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1/22/2007 EECS 723 intro 2/3 Certainly, all electrical engineers know of linear systems theory. But, it is helpful to first review these concepts to make sure that we all understand what this theory is, why it works, and how it is useful. First, we must carefully define a linear-time invariant system. HO: THE LINEAR, TIME-INVARIANT SYSTEM Linear systems theory is useful for microwave engineers because most microwave devices and systems are linear (at least approximately). HO: LINEAR CIRCUIT ELEMENTS The most powerful tool for analyzing linear systems is its eigen function. HO: THE EIGEN FUNCTION OF LINEAR SYSTEMS Complex votages and currents at times cause much head scratching; let’s make sure we know what these complex values and functions physically mean. HO: A COMPLEX REPRESENTATION OF SINUSOIDAL FUNCTIONS Signals may not have the explicit form of an eigen function, but our linear systems theory allows us to (relatively) easily analyze this case as well. Jim Stiles The Univ. of Kansas Dept. of EECS 1/22/2007 EECS 723 intro 3/3 HO: ANALYSIS OF CIRCUITS DRIVEN BY ARBITRARY FUNCTIONS If our linear system is a linear circuit, we can apply basic circuit analysis to determine all its eigen values! HO: THE EIGEN SPECTRUM OF LINEAR CIRCUITS Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 The Linear Time Invariant System.doc 1/13 The Linear, Time- Invariant System Most of the microwave devices and networks that we will study in this course are both linear and time invariant (or approximately so). Let’s make sure that we understand what these terms mean, as linear, time-invariant systems allow us to apply a large and helpful mathematical toolbox! LINEARITY LINEARITY Mathematicians often speak of operators, which is “mathspeak” for any mathematical operation that can be applied to a single element (e.g., value, variable, vector, matrix, or function). ...operators, operators, operators!! For example, a function f ( x ) describes an operation on variable x (i.e., f ( x ) is operator on x ). E.G.: f (y ) = y 2 − 3 g (t ) = 2t y (x ) = x Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 The Linear Time Invariant System.doc 2/13 Moreover, we find that functions can likewise be operated on! For example, integration and differentiation are likewise mathematical operations—operators that operate on functions. E.G.,: d g (t ) ∞ ∫ f ( y ) dy dt ∫ −∞ y ( x ) dx A special and very important class of operators are linear operators. Linear operators are denoted as L [ y ] , where: * L symbolically denotes the mathematical operation; * And y denotes the element (e.g., function, variable, vector) being operated on. A linear operator is any operator that satisfies the following two statements for any and all y : 1. L [ y1 + y2 ] = L [ y1 ] + L [ y2 ] 2. L ⎡a y ⎤ = a L [ y ] , where a is any constant. ⎣ ⎦ Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 The Linear Time Invariant System.doc 3/13 From these two statements we can likewise conclude that a linear operator has the property: L ⎡a y1 + b y2 ⎤ = a L [ y1 ] + b L [ y2 ] ⎣ ⎦ where both a and b are constants. Essentially, a linear operator has the property that any weighted sum of solutions is also a solution! For example, consider the function: L [t ] = g (t ) = 2t At t = 1 : g (t = 1 ) = 2 (1 ) = 2 and at t = 2 : g (t = 2 ) = 2 (2 ) = 4 Now at t = 1 + 2 = 3 we find: g (1 + 2 ) = 2 ( 3 ) =6 =2+4 = g (1 ) + g ( 2 ) Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 The Linear Time Invariant System.doc 4/13 More generally, we find that: g (t1 + t2 ) = 2 (t1 + t2 ) = 2t1 + 2t2 = g (t1 ) + g (t2 ) and g (a t ) = 2 a t = a 2t = a g (t ) Thus, we conclude that the function g (t ) = 2t is indeed a linear function! Now consider this function: y (x ) = m x + b Q: But that’s the equation of a line! That must be a linear function, right? A: I’m not sure—let’s find out! We find that: y ( a x ) = m ( ax ) + b = a mx + b but: a y ( x ) = a (m x + b ) = a mx + a b Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 The Linear Time Invariant System.doc 5/13 therefore: y ( a x ) ≠ a y ( x ) !!! Likewise: y ( x1 + x 2 ) = m ( x1 + x 2 ) + b = m x1 + m x 2 + b but: y ( x1 ) + y ( x 2 ) = ( m x 1 + b ) + ( m x 2 + b ) = m x1 + m x2 + 2b therefore: y ( x1 + x2 ) ≠ y ( x1 ) + y ( x2 ) !!! The equation of a line is not a linear function! Moreover, you can show that the functions: f (y ) = y 2 − 3 y (x ) = x are likewise non-linear. Remember, linear operators need not be functions. Consider the derivative operator, which operates on functions. d d f (x ) dx dx f (x ) Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 The Linear Time Invariant System.doc 6/13 Note that: d d f (x ) d g (x ) ⎡f ( x ) + g ( x )⎤ = + dx ⎣ ⎦ dx dx and also: d d f (x ) ⎡a f ( x )⎤ = a dx ⎣ ⎦ dx We thus can conclude that the derivative operation is a linear operator on function f ( x ) : d f (x ) = L ⎡f ( x )⎤ ⎣ ⎦ dx You can likewise show that the integration operation is likewise a linear operator: ∫ f ( y ) dy = L ⎡f ( y ) ⎤ ⎣ ⎦ But, you will find that operations such as: d g 2 (t ) ∞ ∫ y ( x ) dx dt −∞ are not linear operators (i.e., they are non-linear operators). Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 The Linear Time Invariant System.doc 7/13 We find that most mathematical operations are in fact non- linear! Linear operators are thus form a small subset of all possible mathematical operations. Q: Yikes! If linear operators are so rare, we are we wasting our time learning about them?? A: Two reasons! Reason 1: In electrical engineering, the behavior of most of our fundamental circuit elements are described by linear operators—linear operations are prevalent in circuit analysis! Reason 2: To our great relief, the two characteristics of linear operators allow us to perform these mathematical operations with relative ease! Q: How is performing a linear operation easier than performing a non-linear one?? A: The “secret” lies is the result: L ⎡a y1 + b y2 ⎤ = a L [ y1 ] + b L [ y2 ] ⎣ ⎦ Note here that the linear operation performed on a relatively complex element a y1 + b y2 can be determined immediately from the result of operating on the “simple” elements y1 and y2 . Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 The Linear Time Invariant System.doc 8/13 To see how this might work, let’s consider some arbitrary function of time v (t ) , a function that exists over some finite amount of time T (i.e., v (t ) = 0 for t < 0 and t >T ). Say we wish to perform some linear operation on this function: L ⎡v (t )⎤ = ?? ⎣ ⎦ Depending on the difficulty of the operation L , and/or the complexity of the function v (t ) , directly performing this operation could be very painful (i.e., approaching impossible). Instead, we find that we can often expand a very complex and stressful function in the following way: ∞ v (t ) = a0 ψ 0 (t ) + a1 ψ 1 (t ) + a2 ψ 2 (t ) + = n ∑ an ψ n (t ) =−∞ where the values an are constants (i.e., coefficients), and the functions ψ n (t ) are known as basis functions. For example, we could choose the basis functions: ψ n (t ) = t n for n ≥0 Resulting in a polynomial of variable t: Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 The Linear Time Invariant System.doc 9/13 ∞ v (t ) = a0 + a1 t + a2 t + a3 t + 2 3 = ∑ an t n n =0 This signal expansion is of course know as the Taylor Series expansion. However, there are many other useful expansions (i.e., many other useful basis ψ n (t ) ). * The key thing is that the basis functions ψ n (t ) are independent of the function v (t ) . That is to say, the basis functions are selected by the engineer (i.e., you) doing the analysis. * The set of selected basis functions form what’s known as a basis. With this basis we can analyze the function v (t ) . * The result of this analysis provides the coefficients an of the signal expansion. Thus, the coefficients are directly dependent on the form of function v (t ) (as well as the basis used for the analysis). As a result, the set of coefficients {a1 , a2 , a3 , } completely describe the function v (t ) ! Q: I don’t see why this “expansion” of function of v (t ) is helpful, it just looks like a lot more work to me. A: Consider what happens when we wish to perform a linear operation on this function: Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 The Linear Time Invariant System.doc 10/13 ⎡ ∞ ⎤ ∞ L ⎡v (t )⎤ = L ⎢ ∑ an ψ n (t ) ⎥ = ∑ an L ⎡ψ n (t )⎤ ⎣ ⎦ ⎣ ⎦ ⎣n =−∞ ⎦ n =−∞ Look what happened! Instead of performing the linear operation on the arbitrary and difficult function v (t ) , we can apply the operation to each of the individual basis functions ψ n (t ) . Q: And that’s supposed to be easier?? A: It depends on the linear operation and on the basis functions ψ n (t ) . Hopefully, the operation L [ψ n (t )] is simple and straightforward. Ideally, the solution to L [ψ n (t )] is already known! Q: Oh yeah, like I’m going to get so lucky. I’m sure in all my circuit analysis problems evaluating L [ψ n (t )] will be long, frustrating, and painful. A: Remember, you get to choose the basis over which the function v (t ) is analyzed. A smart engineer will choose a basis for which the operations L [ψ n (t )] are simple and straightforward! Q: But I’m still confused. How do I choose what basis ψ n (t ) to use, and how do I analyze the function v (t ) to determine the coefficients an ?? Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 The Linear Time Invariant System.doc 11/13 A: Perhaps an example would help. Among the most popular basis is this one: ⎧ j ⎛ 2T n ⎞ t ⎜ π ⎟ ⎪e ⎝ ⎠ 0 ≤ t ≤T ⎪ ψn = ⎨ ⎪0 t ≤ 0,t ≥T ⎪ ⎩ and: T T ⎛ 2π n ⎞ 1 1 −j ⎜ ⎟t an = ∫v (t ) ψ n (t ) dt ∗ = ∫v (t ) e ⎝ T ⎠ dt T 0 T 0 So therefore: ∞ ⎛ 2π n ⎞ j⎜ ⎟t v (t ) = n ∑ an e =−∞ ⎝ T ⎠ for 0 ≤ t ≤T The astute among you will recognize this signal expansion as the Fourier Series! Q: Yes, just why is Fourier analysis so prevalent? A: The answer reveals itself when we apply a linear operator to the signal expansion: ⎛ 2π n ⎞ ⎡ ∞ −j ⎜ ⎟t ⎤ ∞ ⎡ − j ⎜ 2T n ⎟ t ⎤ ⎛ π ⎞ L ⎡v (t )⎤ = L ⎢ ∑ an e ⎣ ⎦ ⎝ T ⎠ ⎥ = ∑ an L ⎢e ⎝ ⎠ ⎥ ⎢n =−∞ ⎣ ⎥ n =−∞ ⎦ ⎢ ⎣ ⎥ ⎦ Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 The Linear Time Invariant System.doc 12/13 Note then that we must simply evaluate: ⎡ − j ⎛ 2T n ⎞ t ⎤ ⎜ π ⎟ L ⎢e ⎝ ⎠ ⎥ ⎢ ⎣ ⎥ ⎦ for all n. We will find that performing almost any linear operation L on basis functions of this type to be exceeding simple (more on this later)! TIME INVARIANCE TIME INVARIANCE Q: That’s right! You said that most of the microwave devices that we will study are (approximately) linear, time-invariant devices. What does time invariance mean? A: From the standpoint of a linear operator, it means that that the operation is independent of time—the result does not depend on when the operation is applied. I.E., if: L ⎣x (t )⎤ = y (t ) ⎡ ⎦ then: L ⎡x (t − τ )⎤ = y (t − τ ) ⎣ ⎦ where τ is a delay of any value. Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 The Linear Time Invariant System.doc 13/13 The devices and networks that you are about to study in EECS 723 are in fact fixed and unchanging with respect to time (or at least approximately so). As a result, the mathematical operations that describe most (but not all!) of our circuit devices are both linear and time- invariant operators. We therefore refer to these devices and networks as linear, time-invariant systems. Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 Linear Circuit Elements.doc 1/6 Linear Circuit Elements Most microwave devices can be described or modeled in terms of the three standard circuit elements: 1. RESISTANCE (R) 2. INDUCTANCE (L) 3. CAPACITANCE (C) For the purposes of circuit analysis, each of these three elements are defined in terms of the mathematical relationship between the difference in electric potential v (t ) between the two terminals of the device (i.e., the voltage across the device), and the current i (t ) flowing through the device. We find that for these three circuit elements, the relationship between v (t ) and i (t ) can be expressed as a linear operator! iR (t ) + vR (t ) LR ⎡vR (t )⎤ = iR (t ) = Y⎣ ⎦ R vR (t ) R LR ⎡iR (t )⎤ = vR (t ) = R iR (t ) Z⎣ ⎦ − Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 Linear Circuit Elements.doc 2/6 iC (t ) d vC (t ) + LC ⎡vC (t )⎤ = iC (t ) = C Y⎣ ⎦ dt vC (t ) C t 1 LZ ⎡iC (t )⎤ = vC (t ) = ∫ iC (t ′) dt ′ C ⎣ ⎦ C − −∞ iL (t ) t + 1 LY ⎡v L (t )⎤ = iL (t ) = ∫ vL(t ′) dt ′ L ⎣ ⎦ L −∞ v L (t ) L d iL (t ) LZ ⎡iL (t )⎤ = v L (t ) = L L ⎣ ⎦ dt − Since the circuit behavior of these devices can be expressed with linear operators, these devices are referred to as linear circuit elements. Q: Well, that’s simple enough, but what about an element formed from a composite of these fundamental elements? For example, for example, how are v (t ) and i (t ) related in the circuit below?? Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 Linear Circuit Elements.doc 3/6 i (t ) C + LZ ⎡i (t )⎤ = v (t ) = ??? ⎣ ⎦ L R v (t ) − A: It turns out that any circuit constructed entirely with linear circuit elements is likewise a linear system (i.e., a linear circuit). As a result, we know that that there must be some linear operator that relates v (t ) and i (t ) in your example! LZ ⎡i (t )⎤ = v (t ) ⎣ ⎦ The circuit above provides a good example of a single-port (a.k.a. one-port) network. We can of course construct networks with two or more ports; an example of a two-port network is shown below: i1(t ) i2(t ) C + + v1(t ) L R v2(t ) − − Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 Linear Circuit Elements.doc 4/6 Since this circuit is linear, the relationship between all voltages and currents can likewise be expressed as linear operators, e.g.: L21 ⎡v1 (t )⎤ = v2 (t ) ⎣ ⎦ LZ 21 ⎡i1 (t )⎤ = v2 (t ) ⎣ ⎦ LZ 22 ⎡i2 (t )⎤ = v2 (t ) ⎣ ⎦ Q: Yikes! What would these linear operators for this circuit be? How can we determine them? A: It turns out that linear operators for all linear circuits can all be expressed in precisely the same form! For example, the linear operators of a single-port network are: t v (t ) = LZ ⎡i (t )⎤ = ⎣ ⎦ ∫ g (t − t ′) i (t ′) dt ′ −∞ Z t i (t ) = LY ⎡v (t )⎤ = ⎣ ⎦ ∫ g (t − t ′) v (t ′) dt ′ −∞ Y In other words, the linear operator of linear circuits can always be expressed as a convolution integral—a convolution with a circuit impulse function g (t ) . Q: But just what is this “circuit impulse response”?? Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 Linear Circuit Elements.doc 5/6 A: An impulse response is simply the response of one circuit function (i.e., i (t ) or v (t ) ) due to a specific stimulus by another. That specific stimulus is the impulse function δ (t ) . The impulse function can be defined as: πt ⎞ sin ⎛ ⎜ ⎟ 1 ⎝ τ ⎠ δ (t ) = lim τ →0 τ ⎛πt ⎞ ⎜ ⎟ ⎝ τ ⎠ Such that is has the following two properties: 1. δ (t ) = 0 for t ≠ 0 ∞ 2. ∫ δ (t ) dt −∞ = 1.0 The impulse responses of the one-port example are therefore defined as: gZ (t ) v (t ) i (t ) =δ (t ) and: gY (t ) i (t ) v (t ) =δ (t ) Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 Linear Circuit Elements.doc 6/6 Meaning simply that gZ (t ) is equal to the voltage function v (t ) when the circuit is “thumped” with a impulse current (i.e., i (t ) = δ (t ) ), and gY (t ) is equal to the current i (t ) when the circuit is “thumped” with a impulse voltage (i.e., v (t ) = δ (t ) ). Similarly, the relationship between the input and the output of a two-port network can be expressed as: t v2(t ) = L21 ⎡v1(t )⎤ = ⎣ ⎦ ∫ g (t − t ′) v (t ′) dt ′ −∞ 1 where: g (t ) v2(t ) v (t ) =δ (t ) 1 Note that the circuit impulse response must be causal (nothing can occur at the output until something occurs at the input), so that: g (t ) = 0 for t <0 Q: Yikes! I recall evaluating convolution integrals to be messy, difficult and stressful. Surely there is an easier way to describe linear circuits!?! A: Nope! The convolution integral is all there is. However, we can use our linear systems theory toolbox to greatly simplify the evaluation of a convolution integral! Jim Stiles The Univ. of Kansas Dept. of EECS 1/22/2007 The Eigen Function of Linear Systems 1/7 The Eigen Function of Linear,Time-Invariant Systems Recall that that we can express (expand) a time-limited signal with a weighted summation of basis functions: v (t ) = ∑ an ψ n (t ) n where v (t ) = 0 for t < 0 and t >T . Say now that we convolve this signal with some system impulse function g (t ) : t L ⎡v (t )⎤ = ⎣ ⎦ ∫ g (t − t ′) v (t ′) dt ′ −∞ t = ∫ g (t − t ′) ∑ an ψ n (t ′) dt ′ −∞ n t = ∑ an ∫ g (t − t ′) ψ n (t ′) dt ′ n −∞ Look what happened! Instead of convolving the general function v (t ) , we now find that we must simply convolve with the set of basis functions ψ n (t ) . Jim Stiles The Univ. of Kansas Dept. of EECS 1/22/2007 The Eigen Function of Linear Systems 2/7 Q: Huh? You say we must “simply” convolve the set of basis functions ψ n (t ) . Why would this be any simpler? A: Remember, you get to choose the basis ψ n (t ) . If you’re smart, you’ll choose a set that makes the convolution integral “simple” to perform! Q: But don’t I first need to know the explicit form of g (t ) before I intelligently choose ψ n (t ) ?? A: Not necessarily! The key here is that the convolution integral: t L ⎡ψ n (t )⎤ = ⎣ ⎦ ∫ g (t − t ′) ψ n (t ′) dt ′ −∞ is a linear, time-invariant operator. Because of this, there exists one basis with an astonishing property! These special basis functions are: ⎧e j ωn t for 0 ≤ t ≤T ⎪ ⎛ 2π ⎞ ψ n (t ) = ⎨ where ωn = n ⎜ ⎟ ⎪0 for t < 0,t >T ⎝T ⎠ ⎩ Now, inserting this function (get ready, here comes the astonishing part!) into the convolution integral: Jim Stiles The Univ. of Kansas Dept. of EECS 1/22/2007 The Eigen Function of Linear Systems 3/7 t L ⎡e ∫ g (t − t ′) e dt ′ j ωn t j ωn t ′ ⎣ ⎤= ⎦ −∞ and using the substitution u = t − t ′ , we get: t t −t ∫ g (t − t ′ ) e j ωn t dt ′ = ∫ g (u ) e j ωn (t −u ) ( −du ) −∞ t − ( −∞ ) 0 =e j ωn t ∫ g (u ) e − j ωn u ( −du ) +∞ ∞ =e j ωn t ∫ g (u ) e − j ωn u du 0 See! Doesn’t that astonish! Q: I’m astonished only by how lame you are. How is this result any more “astonishing” than any of the other supposedly “useful” things you’ve been telling us? A: Note that the integration in this result is not a convolution—the integral is simply a value that depends on n (but not time t): ∞ G (ωn ) ∫ g (t ) e − j ωn t dt 0 As a result, convolution with this “special” set of basis functions can always be expressed as: Jim Stiles The Univ. of Kansas Dept. of EECS 1/22/2007 The Eigen Function of Linear Systems 4/7 t ∫ g (t − t ′ ) e j ωn t ′ dt ′ = L ⎡e j ωn t ⎤ = G (ωn ) e j ωnt ⎣ ⎦ −∞ The remarkable thing about this result is that the linear operation on function ψ n (t ) = exp [ j ωnt ] results in precisely the same function of time t (save the complex multiplier G ( ωn ) )! I.E.: L ⎡ψ n (t )⎤ = G ( ωn ) ψ n (t ) ⎣ ⎦ Convolution with ψ n (t ) = exp [ j ωnt ] is accomplished by simply multiplying the function by the complex number G ( ωn ) ! Note this is true regardless of the impulse response g (t ) (the function g (t ) affects the value of G ( ωn ) only)! Q: Big deal! Aren’t there lots of other functions that would satisfy the equation above equation? A: Nope. The only function where this is true is: ψ n (t ) = e j ωn t This function is thus very special. We call this function the eigen function of linear, time-invariant systems. Q: Are you sure that there are no other eigen functions?? Jim Stiles The Univ. of Kansas Dept. of EECS 1/22/2007 The Eigen Function of Linear Systems 5/7 A: Well, sort of. Recall from Euler’s equation that: e j ωn t = cos ωn t + j sin ωn t It can be shown that the sinusoidal functions cos ωn t and sin ωn t are likewise eigen functions of linear, time-invariant systems. The real and imaginary components of eigen function exp [ j ωnt ] are also eigen functions. Q: What about the set of values G ( ωn ) ?? Do they have any significance or importance?? A: Absolutely! Recall the values G ( ωn ) (one for each n) depend on the impulse response of the system (e.g., circuit) only: ∞ G (ωn ) ∫ g (t ) e − j ωn t dt 0 Thus, the set of values G ( ωn ) completely characterizes a linear time-invariant circuit over time 0 ≤ t ≤T . We call the values G ( ωn ) the eigen values of the linear, time-invariant circuit. Jim Stiles The Univ. of Kansas Dept. of EECS 1/22/2007 The Eigen Function of Linear Systems 6/7 Q: OK Poindexter, all eigen stuff this might be interesting if you’re a mathematician, but is it at all useful to us electrical engineers? A: It is unfathomably useful to us electrical engineers! Say a linear, time-invariant circuit is excited (only) by a sinusoidal source (e.g., v s (t ) = cos ωot ). Since the source function is the eigen function of the circuit, we will find that at every point in the circuit, both the current and voltage will have the same functional form. That is, every current and voltage in the circuit will likewise be a perfect sinusoid with frequency ωo !! Of course, the magnitude of the sinusoidal oscillation will be different at different points within the circuit, as will the relative phase. But we know that every current and voltage in the circuit can be precisely expressed as a function of this form: A cos (ωot + ϕ ) Q: Isn’t this pretty obvious? Jim Stiles The Univ. of Kansas Dept. of EECS 1/22/2007 The Eigen Function of Linear Systems 7/7 A: Why should it be? Say our source function was instead a square wave, or triangle wave, or a sawtooth wave. We would find that (generally speaking) nowhere in the circuit would we find another current or voltage that was a perfect square wave (etc.)! In fact, we would find that not only are the current and voltage functions within the circuit different than the source function (e.g. a sawtooth) they are (generally speaking) all different from each other. We find then that a linear circuit will (generally speaking) distort any source function—unless that function is the eigen function (i.e., an sinusoidal function). Thus, using an eigen function as circuit source greatly simplifies our linear circuit analysis problem. All we need to accomplish this is to determine the magnitude A and relative phase ϕ of the resulting (and otherwise identical) sinusoidal function! Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 A Complex Representation of Sinusoidal Functions.doc 1/8 A Complex Representation of Sinusoidal Functions Q: So, you say (for example) if a linear two-port circuit is driven by a sinusoidal source with arbitrary frequency ωo , then the output will be identically sinusoidal, only with a different magnitude and relative phase. C + + v1(t ) =Vm 1 cos (ωot + ϕ1 ) L R v2(t ) =Vm 2 cos (ωot + ϕ2 ) − − How do we determine the unknown magnitude Vm 2 and phase ϕ2 of this output? A: Say the input and output are related by the impulse response g (t ) : t v2(t ) = L ⎡v1(t )⎤ = ⎣ ⎦ ∫ g (t − t ′) v (t ′) dt ′ −∞ 1 We now know that if the input were instead: v1 (t ) = e j ω t 0 Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 A Complex Representation of Sinusoidal Functions.doc 2/8 then: v2(t ) = L ⎡e j ω t ⎤ = G (ω 0 ) e j ω t ⎣ ⎦ 0 0 where: ∞ G (ω 0 ) ∫ g (t ) e − j ω0 t dt 0 Thus, we simply multiply the input v1 (t ) = e j ω t by the complex 0 eigen value G (ω 0 ) to determine the complex output v 2 (t ) : v2(t ) = G (ω 0 ) e j ω t 0 Q: You professors drive me crazy with all this math involving complex (i.e., real and imaginary) voltage functions. In the lab I can only generate and measure real-valued voltages and real-valued voltage functions. Voltage is a real-valued, physical parameter! A: You are quite correct. Voltage is a real-valued parameter, expressing electric potential (in Joules) per unit charge (in Coulombs). Q: So, all your complex formulations and complex eigen values and complex eigen functions may all be sound mathematical abstractions, but aren’t they worthless to us electrical engineers who work in the “real” world (pun intended)? Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 A Complex Representation of Sinusoidal Functions.doc 3/8 A: Absolutely not! Complex analysis actually simplifies our analysis of real-valued voltages and currents in linear circuits (but only for linear circuits!). The key relationship comes from Euler’s Identity: e j ωt = cos ωt + j sin ωt Meaning: Re {e j ωt } = cos ωt Now, consider a complex value C. We of course can write this complex number in terms of it real and imaginary parts: C =a + j b ∴ a = Re {C } and b = Im {C } But, we can also write it in terms of its magnitude C and phase ϕ ! C = C e jϕ where: C = C C ∗ = a 2 + b2 ϕ = tan −1 ⎡b a ⎤ ⎣ ⎦ Thus, the complex function C e j ω t is: 0 Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 A Complex Representation of Sinusoidal Functions.doc 4/8 C e jω t = C e jϕ e jω t 0 0 = C e j ω t +ϕ 0 = C cos (ω 0t + ϕ ) + j C sin (ω 0t + ϕ ) Therefore we find: C cos (ω 0t + ϕ ) = Re {C e j ω t } 0 Now, consider again the real-valued voltage function: v1(t ) =Vm 1 cos (ωt + ϕ1 ) This function is of course sinusoidal with a magnitude Vm 1 and phase ϕ1 . Using what we have learned above, we can likewise express this real function as: v1(t ) =Vm 1 cos (ωt + ϕ1 ) = Re { 1 e j ωt } V where V1 is the complex number: V1 = Vm 1 e j ϕ 1 Q: I see! A real-valued sinusoid has a magnitude and phase, just like complex number. A single complex number (V ) can be used to specify both of the fundamental (real-valued) parameters of our sinusoid (Vm , ϕ ). Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 A Complex Representation of Sinusoidal Functions.doc 5/8 What I don’t see is how this helps us in our circuit analysis. After all: v2(t ) = G (ωo ) (V1 e j ωot ) which means: v2(t ) ≠ G (ωo ) Re { 1 e j ωot } V What then is the real-valued output v2(t ) of our two-port network when the input v1(t ) is the real-valued sinusoid: v1(t ) =Vm 1 cos (ωot + ϕ1 ) ??? = Re { 1 e V j ωot } A: Let’s go back to our original convolution integral: t v2(t ) = ∫ g (t − t ′) v (t ′) dt ′ −∞ 1 If: v1(t ) =Vm 1 cos (ωot + ϕ1 ) = Re { 1 e j ωot } V then: t v2(t ) = ∫ g (t − t ′) Re {V e }dt ′ j ωot ′ 1 −∞ Now, since the impulse function g (t ) is real-valued (this is really important!) it can be shown that: Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 A Complex Representation of Sinusoidal Functions.doc 6/8 t v2(t ) = ∫ g (t − t ′) Re {V e }dt ′ j ωot ′ 1 −∞ ⎧t ⎫ = Re ⎨ ∫ g (t − t ′ )V1 e j ωot dt ′⎬ ′ ⎩ −∞ ⎭ Now, applying what we have previously learned; ⎧t ⎫ v2(t ) = Re ⎨ ∫ g (t − t ′ )V1 e j ωot ′dt ′⎬ ⎩ −∞ ⎭ ⎧ t ⎫ = Re ⎨ 1 ∫ g (t − t ′ ) e j ωot dt ′⎬ V ′ ⎩ −∞ ⎭ = Re { 1 G (ω0 ) e j ωot } V Thus, we finally can conclude the real-valued output v2(t ) due to the real-valued input: v1(t ) =Vm 1 cos (ωot + ϕ1 ) = Re { 1 e j ωot } V is: v2(t ) = Re { 2 e j ωot } V =Vm 2 cos (ωot + ϕ2 ) where: V2 = G (ωo )V1 The really important result here is the last one! Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 A Complex Representation of Sinusoidal Functions.doc 7/8 C + + v1(t ) =Vm 1 cos (ωot + ϕ1 ) L R v2(t ) = Re {G (ωo )V1 e j ωot } − − The magnitude and phase of the output sinusoid (expressed as complex value V2 ) is related to the magnitude and phase of the input sinusoid (expressed as complex value V1 ) by the system eigen value G (ωo ) : V2 = G (ωo ) V1 Therefore we find that really often in electrical engineering, we: 1. Use sinusoidal (i.e., eigen function) sources. 2. Express the voltages and currents created by these sources as complex values (i.e., not as real functions of time)! For example, we might say “ V3 = 2.0 ”, meaning: V3 = 2.0 = 2.0 e j 0 { } ⇒ v3 (t ) = Re 2.0 e j 0e j ωot = 2.0 cos ωot Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 A Complex Representation of Sinusoidal Functions.doc 8/8 Or “ I L = −3.0 ”, meaning: I L = −2.0 = 3.0 e j π ⇒ iL (t ) = Re {3.0 e j π e j ωot } = 3.0 cos (ωot + π ) Or “Vs = j ”, meaning: Vs = j = 1.0 e ( ) j π2 ⇒ v s (t ) = Re 1.0 e { ( ) j ωot j π2 e } ( = 1.0 cos ωot + π 2 ) * Remember, if a linear circuit is excited by a sinusoid (e.g., eigen function exp ⎡ j ω 0t ⎤), then the only unknowns are ⎣ ⎦ the magnitude and phase of the sinusoidal currents and voltages associated with each element of the circuit. * These unknowns are completely described by complex values, as complex values likewise have a magnitude and phase. * We can always “recover” the real-valued voltage or current function by multiplying the complex value by exp ⎡ j ω 0t ⎤ and then taking the real part, but typically we ⎣ ⎦ don’t—after all, no new or unknown information is revealed by this operation! + C + V1 L R V2 = G (ωo )V1 − − Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 Analysis of Circuits Driven by Arbitrary Functions.doc 1/6 Analysis of Circuits Driven by Arbitrary Functions Q: What happens if a linear circuit is excited by some function that is not an “eigen function”? Isn’t limiting our analysis to sinusoids too restrictive? A: Not as restrictive as you might think. Because sinusoidal functions are the eigen-functions of linear, time-invariant systems, they have become fundamental to much of our electrical engineering infrastructure—particularly with regard to communications. For example, every radio and TV station is assigned its very own eigen function (i.e., its own frequency ω )! It is very important that we use eigen functions for electromagnetic communication, otherwise the received signal might look very different from the one that was transmitted! ψ n (t ) ≠ e j ωnt Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 Analysis of Circuits Driven by Arbitrary Functions.doc 2/6 With sinusoidal functions (being eigen functions and all), we know that receive function will have precisely the same form as the one transmitted (albeit quite a bit smaller). Thus, our assumption that a linear circuit is excited by a sinusoidal function is often a very accurate and practical one! Q: Still, we often find a circuit that is not driven by a sinusoidal source. How would we analyze this circuit? A: Recall the property of linear operators: L ⎡a y1 + b y2 ⎤ = a L [ y1 ] + b L [ y2 ] ⎣ ⎦ We now know that we can expand the function: ∞ v (t ) = a0 ψ 0 (t ) + a1 ψ 1 (t ) + a2 ψ 2 (t ) + = n ∑ an ψ n (t ) =−∞ and we found that: ⎡ ∞ ⎤ ∞ L ⎡v (t )⎤ = L ⎢ ∑ an ψ n (t ) ⎥ = ∑ an L ⎡ψ n (t )⎤ ⎣ ⎦ ⎣ ⎦ ⎣n =−∞ ⎦ n =−∞ Finally, we found that any linear operation L [ψ n (t )] is greatly simplified if we choose as our basis function the eigen function of linear systems: Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 Analysis of Circuits Driven by Arbitrary Functions.doc 3/6 ⎧e j ωn t for 0 ≤ t ≤T ⎪ ⎛ 2π ⎞ ψ n (t ) = ⎨ where ωn = n ⎜ ⎟ ⎪0 for t < 0,t >T ⎝T ⎠ ⎩ so that: L ⎡ψ n (t )⎤ = G (ωn ) e j ωnt ⎣ ⎦ Thus, for the example: C + + v1(t ) L R v2(t ) − − We follow these analysis steps: 1. Expand the input function v1 (t ) using the basis functions ψ n (t ) = exp [ j ωnt ] : ∞ v1 (t ) =V01 e j ω 0t +V11 e j ω1t +V21 e j ω2t + = n ∑ Vn =−∞ 1 e j ωnt where: T 1 Vn 1 = ∫ v1 (t ) e − j ωnt dt T 0 Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 Analysis of Circuits Driven by Arbitrary Functions.doc 4/6 2. Evaluate the eigen values of the linear system: ∞ G (ωn ) = ∫ g (t ) e − j ωn t dt 0 3. Perform the linear operaton (the convolution integral) that relates v2 (t ) to v1 (t ) : v2 (t ) = L ⎡v1 (t )⎤ ⎣ ⎦ ⎡ ∞ ⎤ = L ⎢ ∑ Vn 1 e j ωnt ⎥ ⎣n =−∞ ⎦ ∞ = n ∑ Vn =−∞ 1 L ⎡e j ωnt ⎤ ⎣ ⎦ ∞ = n ∑ Vn =−∞ 1 G (ωn ) e j ωnt Summarizing: ∞ v2 (t ) = n ∑ Vn =−∞ 2 e j ωnt where: Vn 2 = G (ωn )Vn 1 and: T 1 ∞ Vn 1 = ∫v 1 (t ) e − j ωnt dt G (ωn ) = ∫ g (t ) e − j ωn t dt T 0 0 Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 Analysis of Circuits Driven by Arbitrary Functions.doc 5/6 C + + ∞ ∞ v1(t ) = ∑ Vn 1 e j ωnt L R v2(t ) = n ∑ G (ωn )Vn =−∞ 1 1 e j ωnt n =−∞ − − As stated earlier, the signal expansion used here is the Fourier Series. Say that the timewidth T of the signal v1 (t ) becomes infinite. In this case we find our analysis becomes: 1 +∞ v2 (t ) = ∫V 2 (ω ) e j ω t d ω 2π −∞ where: V2 (ω ) = G (ω )V1 (ω ) and: +∞ +∞ V1 (ω ) = ∫ v1 (t ) e − j ωt dt G (ω ) = ∫ g (t ) e − jω t dt −∞ −∞ The signal expansion in this case is the Fourier Transform. We find that as T → ∞ the number of discrete system eigen values G (ωn ) become so numerous that they form a continuum—G (ω ) is a continuous function of frequencyω . Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 Analysis of Circuits Driven by Arbitrary Functions.doc 6/6 We thus call the function G (ω ) the eigen spectrum or frequency response of the circuit. Q: You claim that all this fancy mathematics (e.g., eigen functions and eigen values) make analysis of linear systems and circuits much easier, yet to apply these techniques, we must determine the eigen values or eigen spectrum: ∞ +∞ G (ωn ) = ∫ g (t ) e − j ωn t dt G (ω ) = ∫ g (t ) e − jω t dt 0 −∞ Neither of these operations look at all easy. And in addition to performing the integration, we must somehow determine the impulse function g (t ) of the linear system as well ! Just how are we supposed to do that? A: An insightful question! Determining the impulse response g (t ) and then the frequency response G (ω ) does appear to be exceedingly difficult—and for many linear systems it indeed is! However, much to our great relief, we can determine the eigen spectrum G (ω ) of linear circuits without having to perform a difficult integration. In fact, we don’t even need to know the impulse response g (t ) ! Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 The Eigen Spectrum.doc 1/12 The Eigen Spectrum of Linear Circuits Recall the linear operators that define a capacitor: d vC (t ) LC ⎡vC (t )⎤ = iC (t ) = C Y ⎣ ⎦ dt t 1 LZ ⎡iC (t )⎤ = vC (t ) = ∫ iC (t ′) dt ′ C ⎣ ⎦ C −∞ We now know that the eigen function of these linear, time- invariant operators—like all linear, time-invariant operartors—is exp [ j ω t ] . The question now is, what is the eigen spectrum of each of these operators? It is this spectrum that defines the physical behavior of a given capacitor! For vC (t ) = exp [ j ω t ] , we find: iC (t ) = LC ⎡vC (t )⎤ Y ⎣ ⎦ d e j ωt =C dt = ( j ωC ) e j ωt Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 The Eigen Spectrum.doc 2/12 Just as we expected, the eigen function exp [ j ωt ] “survives” the linear operation unscathed—the current function i (t ) has precisely the same form as the voltage function v (t ) = exp [ j ωt ] . The only difference between the current and voltage is the multiplication of the eigen spectrum, denoted as GY (ω ) . C i (t ) = LC ⎡v (t ) = e j ωt ⎤ = GY (ω ) e j ωt Y ⎣ ⎦ C Since we just determined that for this case: i (t ) = ( j ωC ) e j ωt it is evident that the eigen spectrum of the linear operation: d v (t ) i (t ) = LC ⎡v (t )⎤ = C Y ⎣ ⎦ dt is: jπ 2 GY (ω ) = j ω C = ω C e C !!! So for example, if: v (t ) = Vm cos (ωot + ϕ ) {( = Re Vm e jϕ )e }j ωot we will find that: Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 The Eigen Spectrum.doc 3/12 LC ⎡(Vm e j ϕ ) e j ωot ⎤ = GY (ωo ) (Vm e j ϕ ) e j ωot Y ⎣ ⎦ C ( = ωC e jπ 2 ) (V e m jϕ )e j ωot ( = ωC Vm e ( j π 2 +ϕ ) ) e j ωot Therefore: { iC (t ) = Re ωC Vme ( j ϕ +π 2 ) j ωot e } ( = ωC Vm cos ωot + ϕ + π 2 ) = −ωC Vm sin ( ωot + ϕ ) Hopefully, this example again emphasizes that these real- valued sinusoidal functions can be completely expressed in terms of complex values. For example, the complex value: VC = Vme j ϕ means that the magnitude of the sinusoidal voltage is VC =Vm , and its relative phase is ∠VC = ϕ . The complex value: IC = GY (ω )VC C ( = ωC e jπ 2 )V C likewise means that the magnitude of the sinusoidal current is: Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 The Eigen Spectrum.doc 4/12 IC = GY (ω )VC C = GY (ω ) VC C = ωC Vm And the relative phase of the sinusoidal current is: ∠IC = ∠GY (ω ) + ∠VC C =π 2+ϕ We can thus summarize the behavior of a capacitor with the simple complex equation: IC = ( j ωC )VC + IC = ( j ωC )VC VC C ( = ωC e jπ2 )V C − Now let’s return to the second of the two linear operators that describe a capacitor: t 1 vC (t ) = LZ ⎡iC (t )⎤ = ∫ iC (t ′) dt ′ C ⎣ ⎦ C −∞ Now, if the capacitor current is the eigen function iC (t ) = exp [ j ωt ] , we find: Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 The Eigen Spectrum.doc 5/12 t 1 LZ ⎡e C ⎣ j ωt ⎤= ⎦ C ∫e j ωt ′ dt ′ −∞ ⎛ 1 ⎞ j ωt =⎜ ⎟e ⎝ j ωC ⎠ where we assume i (t = −∞ ) = 0 . Thus, we can conclude that: ⎛ 1 ⎞ j ωt LC ⎡e j ωt ⎤ = GZ (ω ) e j ωt = ⎜ Z ⎣ ⎦ C ⎟e ⎝ j ωC ⎠ Hopefully, it is evident that the eigen spectrum of this linear operator is: 1 −j 1 ( j 3π 2 ) GZ (ω ) = C = = e jω C ωC ωC And so: ⎛ ⎞ 1 VC = ⎜ ⎟I ⎝ jωC ⎠ C Q: Wait a second! Isn’t this essentially the same result as the one derived for operator LC ?? Y A: It’s precisely the same! For both operators we find: VC 1 = IC j ω C Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 The Eigen Spectrum.doc 6/12 This should not be surprising, as both operators LC and LC Y Z relate the current through and voltage across the same device (a capacitor). The ratio of complex voltage to complex current is of course referred to as the complex device impedance Z. V Z I An impedance can be determined for any linear, time-invariant one-port network—but only for linear, time-invariant one-port networks! Generally speaking, impedance is a function of frequency. In fact, the impedance of a one-port network is simply the eigen spectrum GZ (ω ) of the linear operator LZ : + I LZ ⎡i (t )⎤ = v (t ) ⎣ ⎦ V =Z I V Z Z = GZ (ω ) − Note that impedance is a complex value that provides us with two things: Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 The Eigen Spectrum.doc 7/12 1. The ratio of the magnitudes of the sinusoidal voltage and current: V Z = I 2. The difference in phase between the sinusoidal voltage and current: ∠Z = ∠ V − ∠ I Q: What about the linear operator: LY ⎣v (t )⎤ = i (t ) ?? ⎡ ⎦ A: Hopefully it is now evident to you that: 1 1 GY (ω ) = = GZ (ω ) Z The inverse of impedance is admittance Y: 1 I Y = Z V Now, returning to the other two linear circuit elements, we find (and you can verify) that for resistors: LR ⎡vR (t )⎤ = iR (t ) Y ⎣ ⎦ ⇒ GY (ω ) = 1 R R LR ⎡iR (t )⎤ = vR (t ) Z ⎣ ⎦ ⇒ GZ (ω ) = R R Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 The Eigen Spectrum.doc 8/12 and for inductors: 1 LL ⎡v L (t )⎤ = iL (t ) Y ⎣ ⎦ ⇒ GY (ω ) = L j ωL LZ ⎡iL (t )⎤ = v L (t ) L ⎣ ⎦ ⇒ GZ (ω ) = j ω L L meaning: 1 1 ( ) j π2 ZR = = R = R ej0 and ZL = = j ωL = ωL e YR YL Now, note that the relationship V Z = I forms a complex “Ohm’s Law” with regard to complex currents and voltages. Additionally, ICBST (It Can Be Shown That) Kirchoff’s Laws are likewise valid for complex currents and voltages: ∑ In n =0 ∑Vn n =0 where of course the summation represents complex addition. Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 The Eigen Spectrum.doc 9/12 As a result, the impedance (i.e., the eigen spectrum) of any one-port device can be determined by simply applying a basic knowledge of linear circuit analysis! Returning to the example: I C + V V L R Z = I − And thus using out basic circuits knowledge, we find: Z = ZC + Z R Z L = 1 j ωC + R j ωL Thus, the eigen spectrum of the linear operator: LZ ⎡i (t )⎤ = v (t ) ⎣ ⎦ For this one-port network is: GZ (ω ) = 1 j ωC + R j ωL Look what we did! We were able to determine GZ (ω ) without explicitly determining impulse response gZ (t ) , or having to perform any integrations! Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 The Eigen Spectrum.doc 10/12 Now, if we actually need to determine the voltage function v (t ) created by some arbitrary current function i (t ) , we integrate: 1 +∞ v (t ) = ∫G (ω ) I (ω ) e j ω t d ω 2π Z −∞ 1 +∞ = 2π ∫( −∞ 1 j ωC + R j ω L ) I (ω ) e j ω t d ω where: +∞ I (ω ) = ∫ i (t ) e − j ωt dt −∞ Otherwise, if our current function is time-harmonic (i.e., sinusoidal with frequency ω ), we can simply relate complex current I and complex voltage V with the equation: V =Z I = ( 1 j ωC + R j ω L ) I Similarly, for our two-port example: C + + V1 L R V2 − − Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 The Eigen Spectrum.doc 11/12 we can likewise determine from basic circuit theory the eigen spectrum of linear operator: L21 ⎡v1 (t )⎤ = v2 (t ) ⎣ ⎦ is: Z L ZR j ωL R G21(ω ) = = ZC + Z L Z R 1 + j ωL R j ωC so that: V2 = G21(ω )V1 or more generally: 1 +∞ v2(t ) = ∫ G21 (ω )V1(ω ) e j ωt d ω 2π −∞ where: +∞ V1(ω ) = ∫ v1(t ) e − j ωt dt −∞ Finally, a few important definitions involving impedance and admittance: Re {Z } Resistance R Im {Z } Reactance X Jim Stiles The Univ. of Kansas Dept. of EECS 1/21/2007 The Eigen Spectrum.doc 12/12 Re {Y } Admittance G Im {Y } Susceptance B Therefore: Z = R + jX Y = G + jB But be careful! Although: 1 1 Y = G + jB = = R + jX Z keep in mind that: 1 1 G ≠ and B≠ R X Jim Stiles The Univ. of Kansas Dept. of EECS