Holography Receiver Design
A. Perfetto and L. D’Addario, 2000-10-06
An overall block diagram of the holography receiver is shown in Figure 1. The signal
path is described in section 2, and then details of the local oscillators are given in section 3.
Design considerations for the horn antennas are described in section 4. The digital assembly,
including correlation and monitor/control, is covered in section 5. Finally, the packaging is
described in section 6.
2. Signal Path
As shown in Figure 1, the signals received by the reference and main antennas are down
converted to a fixed 30 MHz intermediate frequency (IF). The use of a fixed IF frequency should
reduce phase errors in the IF system.
Two Millitech biasable balanced mixers and a common local oscillator (LO) are used for
the signal first down conversion. The mixers are optimized for operation at 79 and 104 GHz,
have a typical conversion loss of 11 dB and require about 1mW of LO power. Specifications for
the mixers are given in Appendix B.
The two IF signals are first amplified by low noise figure (1.3 dB) bipolar amplifiers,
yielding an estimated SSB noise temperature of about 3900K. The signals are then band limited
using a 1.5 MHz band pass filter centered at 30 MHz. The IF levels of the channels will be
different; on boresight, the signal channel level is about 40 dB higher than the reference channel.
Two programmable signal attenuators, one for each channel, are used to set the relative
level between channels, prevent saturation of the second 30 MHz IF amplifier and set the
appropriate input level to the SSB mixer. The attenuators have a 63 dB range in 1 dB steps and
require 6 bits for attenuation control.
The receiver’s predicted interstage signal levels are shown in Figure 3. The power levels
shown assume a 1uW photodetector output power, 30 dB gain for the transmitter and reference
feed horns and IF attenuator set at 0 dB.
Both 30 MHz main and reference signals are further down converted to a baseband signal
and a SSB mixer is used to select the upper sideband in this final mixing process to 15 KHz. The
required IF to baseband gain is about 45dB.
The two baseband signals are filtered using an anti-aliasing 8 KHz band pass filter (10
KHz at -3db points). The main signal is divided in two and one half is shifted 90o in phase by a
broad band phase shifter resulting in the S and Q signals. S is the instantaneous voltage
(proportional to field intensity), Q is this voltage shifted by 90o.
The reference channel is also divided in two, but only the instantaneous voltage R is
required for correlation. The 15 KHz S, R and Q output signals are amplified to 5Vp-p nominal
and connected to the DSP card where the signal are digitized and correlated and sent to the AMB
interface (CAN bus).
Page 1 of 9
2.1 IF Chain Control and Monitor
Control: Signal Attenuator 6b
Reference Attenuator 6b
Monitor: 30 MHz signal monitor 0-5 V
30 MHz reference monitor 0-5 V
15 KHz signal monitor 0-5 V
15 KHz reference monitor 0-5 V
3. Local Oscillators
To minimize the number of oscillators required for the holography receiver, the input
signals are down converted from mm-wave to baseband with just two frequency conversions. A
phase locked Gunn oscillator system is used as LO for the first downconverter and a high
stability oven-controlled crystal oscillator (OCXO) as the LO for the second downconverter.
3.1 First LO system
A block diagram of the holography receiver first local oscillator is shown in Figure 2.
The LO signal for the millimeter wave mixers is generated by two varactor tuned Gunn
oscillators, one operating at 78.920 GHz (LOL) and the other at 104.020 GHz (LOH). Both have
an electrical tuning range of +-150 MHz.
The choice of employing two oscillators instead of one single mechanically tuned unit
was based on the difficulty of purchasing oscillators with the required wide tuning frequency
band and output power. Also, remotely servo-controlling two mechanical micrometers (back-
shorts for frequency tuning and power peaking) would add an extra complexity to the system.
The specifications for the varactor tuned Gunn oscillators are given in Appendix B.
As is shown in Fig. 2, just one Gunn oscillator can be turned on at a time. The selection
of oscillators is accomplished by switching the Gunn diode and varactor bias to the desired
oscillator with a Gunn_Select bit from the MC card. The oscillator will be automatically selected
when the desired holography frequency is entered in the control system.
Power levels shown in Fig. 2 are estimates based on typical components insertion losses
and with the mechanical attenuators set at their minimum value. All millimeter wave components
are in the 75-110 GHz waveguide band. Oscillator output power is monitored at the power
detector, amplified and sent to the MC card.
The oscillator power is split in the matched hybrid tee and fed to the reference and signal
mixers. There is enough isolation between the mixers to prevent any channel crosstalk problems.
There are two important frequency related requirements for the first local oscillator: a)
Capability to tune to the transmitter frequency and tuning step size and b) Automatic Gunn
oscillator phase-lock acquisition.
The Gunn oscillator frequency is phase-locked to a high stability 95 MHz OCXO
employing conventional heterodyne technique. A sample of the Gunn 78.89 or 103.98 GHz is
downconverted to a Lock-IF near 95 MHz by mixing it with a harmonic N (N=9,11) of the output
frequency of a 8.6 - 11 GHz frequency synthesizer (FSYN) which is phase locked to a 25 MHz
reference. The Lock-IF and Fsynt paths are separated in the triplexer.
Page 2 of 9
The Lock-IF signal is connected to a digital frequency/phase detector where its frequency
and phase are compared with those of the 95 MHz (FOFFSET) OCXO. The detector’s output is an
error signal proportional to the frequency and phase difference between the Lock-IF and FOFFSET.
Frequency and phase lock acquisition is achieved by applying this error signal to the Gunn’s
varactor bias tuning. The frequency locking range is approx. +- 100 MHz. If the Lock-IF
frequency is within this range the receiver LO will acquire lock automatically.
The frequency/phase detector circuitry design is the same used in NRAO mm-wave
receivers and it will only require modification for the varactor bias voltage range.
The holography frequency tuning scheme is designed to give a fixed IF for any
transmitter signal frequency. This is done by making the receiver frequency synthesis scheme
similar to that of the transmitter:
fXMTR = N fSYN1 + fOFFSET1
fLO = N fSYN2 + fOFFSET2
For the same harmonic number N:
fIF = fXMTR - fLO (USB)
= N ∆fSYN + ∆fOFFSET
= N (i δf1 + j δf2) + ∆fOFFSET
where δf1 and δf2 are the resolution of the synthesizers.
Let δf1 = δf2 = δf and fOFFSET1 = 125 MHz (fixed). We choose fOFFSET2 so that fIF =
∆fOFFSET1 giving a fixed IF for any available signal frequency. In our LO scheme, fIF = 30 MHz
and fOFFSET2 = 95 MHz.
The synthesizer is a Micro Lambdas model MLSL-0811. It has a frequency resolution of
125 KHz and phase noise of -60 dBc/Hz @ 1KHz offset. The synthesizer’s frequency can be set
by commands that are sent to the MLSL synthesizer via a three-wire bus consisting of a Select
line, a Data line, and a Clock line. The 3-wire bus will be driven by the DSP (see section 5
The 95.0 MHz offset oscillator is an free running OCXO from Temex with frequency vs
temperature change of <2E-8 and long term stability of < +-1E10-7/year. It requires 12V @
500mA at warm-up and 120mA at 25o C.
3.2 Second Local oscillator
The receiver’s second local oscillator is also a free running 29.85 MHz OCXO from
Temex with similar frequency stability and power requirements. Its output power is
3.3 LO Control and Monitor
The required monitor and control parameters are:
Page 3 of 9
LO status: Gunn Oscillator Lock Alarm 1b
Synthesizer Lock Alarm 1b
Control: Gunn Oscillator On/Off 1b
Gunn Oscillator Select 1b
PLL open 1b
Synthesizer Frequency Serial Data
Monitor: LO level Analog
LO plate temperature Analog
4. Horn Antennas
The receiver will incorporate two horn antennas, mounted on opposite ends of the same
box at the apex of the 12 m antenna. One horn is a prime focus feed for the antenna, and the
other (pointing outward) receives the reference signal directly from the transmitter. In addition,
the transmitter will use a horn antenna. Each horn will be designed and fabricated specifically
for this application. Each has different requirements, summarized in the following table.
The transmitter's beamwidth is determined by the need to illuminate the 12 m antenna
with reasonable uniformity. Variations in phase, if uncorrected, lead directly to errors in the
reflector surface deviations being measured. It is expected that the horn can be made to produce
a far-field spherical wave to the required tolerance within the part of the beam being used, but
this will be checked by careful measurement on a test range (see below).
The reference antenna is the least critical. Its gain in the direction of the transmitter
should remain high over the scan angle range. The complex gain pattern over this range, to the
extent that it is not constant, has a small effect on the transverse resolution of the final surface
deviation map because its Fourier transform is convolved with the ideal point spread function
determined by the sampling of the beam map.
The transmitter and reference horns will have gains of 34 dB if the beamwidths are as
specified. A conservative value of 30 dB is used in the power budget calculations.
It can be seen that the transmitter and reference horn beamwidths are the same, so the
same design will be used for both. A circular, corrugated, narrow-angle horn is appropriate, in
view of the circularity and frequency range requirements. A lens-corrected aperture is necessary
in order to keep the length from being impractically large. A broad-band design covering the
75-110 GHz waveguide band will be used. The 0.5 dB beamwidth will vary somewhat over the
band, but should remain within 5% and this should be satisfactory.
The signal feed is the most critical horn for this application. The f/0.4 reflector subtends
an angle of 128 deg at the prime focus. The feed needs to illuminate it with reasonably uniform
amplitude; we choose -3 dB edge taper. The critical thing is the phase pattern, which directly
affects the surface deviation measurements. It does not seem feasible to achieve uniform phase
within 0.6 deg over such a large angle, so we will rely on careful measurement of the feed on a
test range. The facilities at IRAM (Grenoble) will be used for this measurement. Similar
measurements will be made on the other horns. A wide-flare-angle, corrugated, circular horn
(scalar feed) is appropriate. It will probably be possible to scale an existing design to our
Page 4 of 9
There is considerable experience within the NRAO in the design and fabrication of
corrugated horns. Although the detailed designs for this project have not yet been done, we
anticipate no difficulty in meeting the requirements.
Horn Antenna Requirements
Frequencies 78.9 and 104.0 GHz
Weatherproofing Lens or membrane across aperture
Dry air pressurization
Beamwidth, -0.5 dB 2.3 deg (subtended by 12m antenna)
-3.0 db 4.6 deg nominal
Polarization, nominal vertical
cross <-30 dB within 1.2 deg
Beam circularity <0.5 dB within 1.2 deg
Phase uniformity <0.3 deg within 1.2 deg (2.5 microns)
Beamwidth, -0.5 dB 2.3 deg (scan angle)
-3.0 dB 4.6 deg nominal
Polarization, nominal vertical
cross <-30 dB within 1.2 deg
Beam circularity <0.5 dB within 1.2 deg
Beamwidth, -3.0 dB 128 deg (edge of 12 m dish)
Polarization, nominal vertical
cross <-20 dB within 64 deg of center
Uniformity, amplitude <0.2 dB from best-fit circular
gaussian within 64 deg
Uniformity, phase <5 deg from constant within 64 deg
Knowledge, phase <0.6 deg within 64 deg (5 microns)
(includes measurement accuracy and stability)
Page 5 of 9
5. Digital Assembly
The digital assembly provides analog to digital conversion for the baseband signals from
the receiver, performs all auto- and cross-correlations, and makes the correlation results available
for reading by the monitor-control system over the AMB (CAN bus). It also handles control and
monitor data between MC and the receiver. Timing of the data sampling and of the integration is
determined within this assembly, based on system-synchronous references at 20.833 Hz and 25
A block diagram of the assembly is shown in Figure 4. The main signal path is from the
baseband inputs through three 16-bit ADCs to a digital signal processor (DSP), where the
correlations are performed, and then via a serial link to the AMB interface. At low priority, the
DSP also acts as host to the microwave synthesizer, which requires serial data for control. The
AMB interface uses the AMBSI board described in , with firmware specific to the holography
receiver. It supports latched control bits to the receiver, as well as input of analog and digital
monitor signals. The logical interface over the AMB is described in detail in .
5.2 Dynamic Range
The ADCs are a significant factor in determining the dynamic range of the receiver. We
calculate the required range as follows. From , the minimum transmitter power is 9\micro W
EIRP when the receiver system temperature is 3200K. The ratio of these, which determines the
on-boresight SNR via the geometry and antenna gains, is fixed by the desired surface
measurement accuracy. If we process a 10 kHz bandwidth with Nyquist rate sampling, the noise
power at the receiver input is 4.4e-16 W and the on-boresight signal power is 9e-10 W, for an
SNR of 63 dB. It is likely that we will achieve a much higher SNR because the transmitter
power will greatly exceed the minimum required and the system temperature may be lower.
Allowing for 20 dB additional SNR, we have a maximum SNR of 83 dB (power) or 1.4e4 in
voltage, which corresponds to 14 bits. The ADCs selected (Analog Devices type AD976) have
16 b of resolution, and in addition are specified to have a spurious-free dynamic range of 100 dB.
5.3 Data Rates and Processing Speed
At a sampling rate of 20 kSps, the total input rate for the three signals is 60000 16b words
per second. To perform all six correlations (three self-products and three cross-products) then
requires 120000 multiply-accumulate (MAC) operations per second. In most DSPs, a MAC is
performed in a single instruction. If we allow another instruction per signal sample for reading
the ADCs and yet another per sample period for triggering all three conversions, we get a total
rate of 200 kips for the basic processing. This is easily handled by many kinds of
microprocessor, including DSPs from at least three manufacturers (Motorola, Texas Instruments,
Analog Devices). A fixed-point processor with 16b word length and 10 MHz clock is more than
Several factors may require a much higher instruction rate. First, higher than 16b
precision is needed in the MACs to avoid overflow. Some processors provide this inherently,
otherwise double-precision calculations would need to be programmed; this would
approximately double the instruction rate. Second, we may wish to operate at higher than
Nyquist sampling rate as a contingency against various difficulties, like achieving sufficient
Page 6 of 9
tuning resolution or transmitter line width. Third, we may wish to try implementing the 90
degree phase shift of the Q channel digitally (Hilbert transform filter); this requires at least 2x
oversampling as well as additional processing. Finally, some processor instructions are needed
to handle the data transfer to the AMB interface, as well as to handle control of the microwave
synthesizer; however, these last functions will produce an extremely small load compared with
the main signal processing.
Data transfer to the AMB interface will involve 6 32-bit words for each integration. At
the rate of one integration every 12 msec, this is an average of 692 bytes/sec. Such a rate is
easily handled by an SPI interface, which is the planned method. Actual transfers will be in
higher-speed bursts, once per 48 msec timing interval. This is 96 bytes per timing interval. The
AMB interface then transfers the data to the Antenna Bus Master computer using the CAN bus.
Each CAN transaction can transfer 8 data bytes, so 12 transactions are needed per timing
interval. The bus capacity is around 50 transactions per timing interval for system-synchronized
monitor data. Although this is a significant load, it is easily handled because very few devices
are active on the bus during holography observations. The only other significant load in this
mode is the antenna control unit.
Efficient organization of the DSP firmware implies that the input data should be buffered
for an entire integration before being correlated. This is because no DSPs have six MACs that
can be operated in parallel. It is thus better to compute each of the six correlations completely
before going on to do the next. If the maximum integration time is 48 msec, then 2880 words
(16b) are needed for the buffer. Much larger on-chip data memories are available on most DSPs,
so that longer integration times can probably be accomodated within this structure.
Based on these considerations, the Analog Devices ADSP-2185 has been identified as
suitable. It is capable of 33 Mips, but will be operated at 25 Mips (see Timing, below); it has
32kwords of on-chip RAM. Many other choices are possible, and the final selection will be
based on practical considerations such as the availability and cost of development tools. There is
experience in the group with other Analog Devices DSPs of the same family. In view of the
relatively low complexity of the code, programming in assembly language is efficient for coding
and documentation, as well as for execution; however, any available high level languages will be
considered during the detailed design.
It is intended to operate the DSP and ADCs at higher than Nyquist rate to the extent that
processing speed permits.
The 20.833 Hz (48 msec) system timing signal is brought directly to the DSP chip, where
it generates an interrupt. This triggers the start of an integration cycle, consisting of four
12-msec integrations. An internal timer in the DSP is used to start each integration after the first.
The DSP clock is taken from the 25 MHz system reference, rather than from a free-running
crystal. This ensures that any time interval generated within it is exactly known and
synchronous. The sampling clock (nominally 20 kHz) will also be generated by the DSP and will
be synchronous with the system.
It should be possible to make successive integrations contiguous, with no intervening
blanking time. However, should such blanking be desired it can easily be programmed with a
resolution of one sampling interval (1/20kHz = 50 \micro sec).
Page 7 of 9
5.5 Other Circuit Details
A small PLD chip will be used for miscellaneous "glue" logic for control of the ADCs.
The ADSP-2185 includes two hardware serial ports. One will be programmed as a
bi-directional SPI for communication with the AMB interface. The other will be programmed to
control the microwave synthesizer. (This may be done differently if another DSP chip is selected
during detailed design.)
The receiver requires 16 control bits, consisting of two 6-bit IF attenuator settings (one
each for signal and reference channels), 1b to turn the Gunn oscillator on or off, 1b to the select
either the low- or high-frequency Gunn oscillator, 1b to force the PLL open for testing, and 1
spare bit. Two status bits, indicating phase locking of the microwave synthesizer and of the
Gunn oscillator, are returned; and two spare status bits are provided. Ten analog monitor points
are supported at 10b resolution, including 5 signal levels, harmonic mixer current, Gunn
oscillator tuning voltage, temperature control heater current, and 2 temperatures; 6 spare analog
monitors are also available. All of this is handled by the AMBSI board.
The assembly thus consists of the AMBSI board (existing hardware design, additional
firmware needed) and one other PCB containing the remaining circuits (to be designed). Each of
the boards will be smaller than 150 cm^2. Total power consumption is expected to be less than
5W, using only a single d.c. supply at 5V.
5.6 Alternatives Considered
The data processing rate for correlation is far lower than is provided by even "slow"
DSPs. The correlation could also have been implemented in hardware, e.g. by using a small
FPGA. The NRAO has considerable experience and development tools for the Xilinx family of
FPGAs, so this would be straightforward. Still, a small microprocessor would be necessary to
manage the data transfer and to perform the auxiliary control and monitoring functions. For this
reason, and to maintain maximum flexibility for future uses, the DSP solution was selected. The
total chip count is probably smaller, and there is no significant effect on cost.
A more extreme option would be to move even more of the processing into the digital
domain by digitizing at the ~30 MHz IF. This is feasible, but then the necessary digital filters
and correlators would have to be implemented in hardware. A large and fast FPGA would
probably be able to accomplish this. It was our judgment that this implementation would be less
straightforward and therefore would pose considerable risk to the schedule than the one planned.
The cost saving in RF hardware (eliminating the second downconversion) would be offset by
having to use faster and more complex digital devices and by increased manpower for design and
debugging. No performance improvement could be identified. For these reasons, this approach
was not adopted.
All receiver electronics (except power supplies) are packaged in a compact, temperature
stabilized box. The power supplies are inside an auxiliary chassis mounted in the antenna’s
receiver cabin and the DC power is sent to voltage regulators in the receiver using a multi-
To maintain phase and gain stability all critical components will be mounted on a
thermally regulated base plate. The power dissipation in the receiver is about 35 watts.
Page 8 of 9
The receiver box size is designed to fit inside the apex trough hole (375 mm dia) and it
will be installed on the translation stages mounting plate. The requirement of focusing the
receiver to both near and far field positions is met by choosing between two receiver’s box
mounting flanges. The package relative small size will facilitate the installation process. The
details of the mechanical interface are given in Figure 5 “Holography/Apex Mechanical
Receiver physical dimensions and electrical interface requirements are given in the table
Size: 250 x 250 x 620 mm
Mass: Approx. 25 Kg (includes mounting flange)
Cables required to run receiver:
25 MHz Reference 1 RG-214
Monitor & Control 1 AMB
DC Power Supplies 1 Multi-conductor (TBD)
Cables run from apex to the antenna’s receiver cabin
 M. Brooks, "ALMA Monitor and Control Bus Draft Specifications," Version 1.3,
 M. Pokorny, "Holography Receiver/Monitor Control Interface," ICD
ALMA09003.08000.0001, draft dated 2000-Oct-02.
 "ALMA Holography System Overview," NRAO, document prepared for holography system
Page 9 of 9
RF 104.020 GHz (+/-150MHz)
H 30 MHz SIGNAL
TEST 15KHz SIGNAL TEST
RF L 78.920 GHz (+/-150MHz)
30/1.5 MHz 63/1dB 15 KHz (USB)
SSB S SIGNAL CHANNEL
MIXER BPF 15 KHz OUTPUTS
90 Q TO A/D PCB
G=19dB G=24dB 8 KHz
0 dBm 6 DETEC. DET
SSB CONVERTER AND
30 MHz SIG MONITOR
30 MHz SIGNAL LO2
LEVEL ADJUST (TO M/C PCB)
(TO M/C PCB)
(TO M/C PCB)
(20mW min) LO H
-10 -10 0 dBm
HARM MXR VARACTOR TUNED
L GUNN OSCILLATORS 25 MHz
78.890 GHz STANDARD (FROM ANTEN.
PLL (0 dBm)
MONITOR GUNN OSCIL.
(TO M/C) PHASE-LOCK
30 MHz REFER.
TO M/C PCB TEST LO2 15KHz REF. TEST
30/1.5 MHz 63/1dB 15 KHz (USB)
R REF. CHANNEL
BPF 15 KHz OUTPUT
TO A/D PCB
G=30dB G=56dB 8 KHz
NF=1.3 (NOT USED)
SSB CONVERTER AND DET
BIASABLE 30 MHz REF. MONITOR
BALANCED MIXER 30 MHz REF. LO2
LEVEL ADJUST (TO M/C PCB)
(TO M/C PCB)
(TO M/C PCB)
15 KHz A/D CONVERTERS CAN TO ALL VOLTAGE
INPUTS BUS REGULATORS
R DC POWER CABLE
20.833 Hz (FROM B.E.
TO M/C TEMPERATURE
CAN INTERFACE 25 MHz REF. PCB
Fig. 1 - Holography Receiver Front-end
0 dBm TEE 0 dBm
RFH = 104020 MHz
-1.5 -1.5 -3.5 -1.5 -1.5
RFL = 78920 MHz
IF = 30 MHz
ISOLATOR LO H = 103980 MHz = FSYN . N + 95 MHz
30 MHz I.F. 30 MHz I.F. FOR N = 11 FSYN = 9445 MHz
REF CH SIGNAL CH MIN. LO FREQ. STEP SIZE = 1.375 MHz
(TO M/C) LO L = 78890 MHz = FSYN . N + 95 MHz
FOR N = 9 FSYN = 8755 MHz
FSYN MIN. LO FREQ. STEP SIZE = 1.125 MHz
10dB -1.5 LO H AND LO L TUNING BW IS +- 150 MHz OF Fc
-15 dBm N N = HARMONIC NUMBER
+13 dBm (min)
Varactor Tuned LOCK ALARM
LO H LO L PHASE/FREQUENCY VARAC. VOLT. MON.
DETECTOR HARM MXR I_MON
GUUN BIAS VOLTAGE +13 dBm
0 TO 20V LOCK
VOLTAGE 95.000 MHz 8.6 - 11.0 ALARM
SW-2 Oven Controlled SYNTHESIZER
+10 V SW-1
Crystal Oscill. 125KHz RESOL.
GUNN ON/OFF GUNN_SELECT (SERIAL PORT)
(DIG BIT) (DIG BIT) (TO M/C)
(TO M/C) (TO M/C)
Fig. 2 - Holography Receiver 1st LO Diagram
DATA TO/FROM RECEIVER HARDWARE
SYNTHESIZER ANALOG MONITOR SIGS CONTROL BITS STATUS BITS
serial data 0-5 V 6b signal attenuator 1b synth lock
set frequency 4x signal levels 6b ref attenuator 1b GDO PLL lock
1x LO level 1b GDO on/-off 2b spare
synthset 2x LO internals 1b GDO select hi/low
1x heater current 1b PLL open
2x temperature 1b spare
analogin controlbits monbits
16 chan, 10b
AIN D0..15 D0..15 SP1
16b ADC FO SP2 SPI AMB
RD DSP RESET
R INTERFACE AMB (CAN)
RECEIVER AIN D0..15 ADSP-2185 INTERFACE
OUTPUTS 16b ADC SIGNALS
(15 kHz) CS
20.8 Hz REF
25 MHz REF
HOLOGRAPHY RECEIVER -- DIGITAL ASSEMBLY
Size Document Number Rev
Figure 4: Digital Assembly A ALMA09003S0001
Date: Wednesday, October 04, 2000 Sheet 1 of 1
4 3 2 1
RCVR MOUNT FOR
NEAR FIELD POSITION ZONE REV DESCRIPTION DATE APPROVED
RCVR MOUNT FOR
FAR FIELD POSITION
D CONNECTORS (3EA) FEED HORN D
HOLOGRAPHY RECEIVER PHYSICAL DIMENSIONS AND REQUIREMENTS:
1. SIZE: 250 x 250 x 620 mm
2. MASS: APPROX. 25 Kg
620 3. RECEIVER WILL BE INSTALLED ON THE SUB-REFLECTOR'S
TRANSLATION STAGES MOUNTING FLANGE.
4. NEAR AND FAR FIELD FOCAL POSITIONS ARE SET BY CHANGING
FOCUS STAGE RECEIVER MOUNTING FLANGES.
5. CABLES REQUIRED TO RUN RECEIVER:
STAGE TRAVEL: +/- 15 mm 90 95 RX MOUNTING PLATES 25MHz REF. CABLE: 1 (RG-214)
9.5 mm THICK
NEAR FIELD PRIMARY FOCAL MONITOR & CONTROL CABLE: 1 (AMB)
C FOCAL POINT
PRIMARY FOCAL SPACERS (4EA)
POINT DC POWER SUPPLIES CABLE: 1 (MULTI-CONDUCTOR) C
Ø375 CABLES RUN FROM APEX TO ANTENNA RECEIVER CABIN.
NOTE: RELATED DOCUMENT AND DRAWING: ALMA-US ICD No. 2
ANTENNA/APEX INTERFACE DWG # 03020810M002A
B Ø500 Ø500 B
LOOKING INTO HOLOGRAPHY PART OR NOMENCLATURE MATERIAL FIND
CAGE CODE SYM
QTY REQD IDENTIFYING NO. OR DESCRIPTION SPECIFICATION NO.
MAIN FEED HORN
UNLESS OTHERWISE SPECIFIED CONTRACT NO.
DIMENSIONS ARE IN MILLIMETERS NATIONAL RADIO ASTRONOMY OBSERVATORY
operated by ASSOCIATED UNIVERSITIES INC. in agreement with
DECIMALS ANGLES THE NATIONAL SCIENCE FOUNDATION
METRIC DO NOT SCALE DRAWING DRAWN
A.A.Perfetto 2000-10-03 Holography/Apex
A THIRD ANGLE PROJECTION
Mechanical Interface A
SIZE CAGE CODE DWG NO. REV.
SIMILAR TO SPECIAL MARKING SYM
SCALE CALC. WT ACT. WT SHEET
4 3 2 1
APPENDIX 'A' - HOLOGRAPHY RECEIVER PARTS LISTS AND COSTS
Item Description Supplier Model # Qty Cost each Cost Qty Delivery Basis
Millimeter-wave Compontes :
1 Biasable Balanced Mixers - 78 - 105 GHz Millitech MXB-10-RR0WF 3 $2,475 $7,425.00 8w quote
2 Gunn Oscillator - 104 GHz Spacek Labs GW-103 2 $2,590 $5,180.00 8w quote
3 Gunn Oscillator - 78.9 GHz Spacek Labs GW-789 2 $2,590 $5,180.00 8w quote
4 Reference Feed Horn TBD N/A 2 $1,000 $2,000.00 Feb/01 estimate
5 Main Feed Horn TBD N/A 2 $1,500 $3,000.00 Feb/01 estimate
6 Matched Hybrid Tee Aerowave 10 2911 2 $1,400 $2,800.00 4w catalog
7 10 dB Coupler Aerowave 10 3000 2 $950 $1,900.00 4w catalog
8 Harmonic Mixer Pacific Millimeter WM 2 $640 $1,280.00 4w catalog
9 Detector Pacific Millimeter WD 2 $770 $1,540.00 4w catalog
10 Isolator Quinstar QIF-1000-AA 4 $890 $3,560.00 6w quote
11 Attenuator Aerowave 10 2220 2 $725 $1,450.00 8w catalog
12 75-115 GHz Miscel. wavequide Aerowave 10-xxxx 1 $1,000 $1,000.00 4w catalog
RF and BB Components:
13 Phase Locked Freq. Synthesizer 8.6 - 10.5 GHz Micro Lambda MLSL-xxxx 2 $3,200 $6,400.00 8w quote
14 Oven Controlled Crystal Oscillator - 95.000 MHz Wenzel 500 08181 2 $612 $1,224.00 10w quote
15 Oven Controlled Crystal Oscillator - 29.985 MHz Temex QEO-67-CO-... 2 $650 $1,300.00 10w quote
16 Digital Frequency/Phase Detector Box NRAO N/A 2 $1,000 $2,000.00 Mar/01 estimate
17 Triplexer Pacific Millimeter MD2A 1 $220 $220.00 4w quote
18 RF Amplifier +19dB Miteq AU-1A-0520 1 $290 $290.00 8w
19 RF Amplifier +30dB Miteq AU-2A-0120 1 $300 $300.00 8w
20 RF Amplifier +24dB Miteq AU-1421 1 $300 $300.00 8w
21 RF Amplifier +56dB Miteq AU-1494 1 $350 $350.00 8w
22 Band Pass Filter 30MHz / 1.5 MHz BW Reactel AB6-30-1.5S11 2 $400 $800.00 8w quote
23 Programmable Step Attenuator (63 dB /1dB) Weinschel 3230 2 $500 $1,000.00 8w quote
24 3-Way Power Splitter Mini-Circuits ZFSC-3-13 2 $52 $104.00 2w catalog
25 2-Way Power Splitter Mini-Circuits ZFSC-2-6 2 $50 $100.00 2w catalog
26 Miscel. RF (e.g. connectors, pads) N/A N/A 1 $500.00 $500.00 Mar/01 estimate
27 SSB Mixer and Quadrature Box NRAO NRAO 2
N/A 3 $700 $2,100.00 Mar/01 estimate
28 DSP Card NRAO N/A 2 $500 $1,000.00 Mar/01 estimate
29 Voltage Regulators Box NRAO N/A 2 $300 $600.00 Mar/01 estimate
30 Receiver Enclosure and Mounting Plates Machine Shop N/A 1 $4,000 $4,000.00 Mar/01 estimate
31 Power Supplies/Eclosure/RF and Power Cables TBD N/A 1 $2,000 $2,000.00 Mar/01 estimate
Note: Machine Shop estimates: $50/hour TOTAL: $60,903.00
APPENDIX ‘B’ - MM-WAVE MIXERS AND GDO SPECIFICATIONS
Biasable Balanced Mixer RF 104 GHz, 78.9 GHz
Millitech IF 30 MHz
MXB-10-RR0WF LO 103.97 GHz, 78.875 GHz
LO Power 1 mW
NF or Conv. Loss 11 dB (max) (SSB)
Bias Voltage +15V
Varactor Tuned Gunn Oscillator Center Frequency 103.98 GHz
Spacek Labs Output Power 30 mW min
GW-103 Tuning Bandwidth + - 150 MHz
Varactor Voltage 0 - 20 volts
Gunn Bias +10 volts
Varactor Tuned Gunn Oscillator Center Frequency 78.890 GHz
Spacek Labs Output Power 50 mW min
GW-789 Tuning Bandwidth + - 150 MHz
Varactor Voltage 0 to + 20 volts
Gunn Bias +10 volts