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# Communication options for Command and Control

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```									Direct Current Measurement:
a NEPTUNE Power System white paper

Harold Kirkham

1.      Scope
This document addresses the measurement of direct current for NEPTUNE. The purpose is to
review the methods that might be used for control and for protective relaying, and make recom-
mendations for consideration by the Power Group.
Definition
There are typically two aspects to the problem of current measurement: transduction either into
some useful scale or into some other quantity, and isolation from the circuit being measured. For
this white paper, current measurement means the obtaining of a value at one location of the
current at another, with suitable isolation from the voltages involved. Thus, it is specifically
intended that the isolation function be included in these considerations.

2.      Review of available current measurement methods
A Taxonomy
Consider first the transduction aspects of the measurement of current. A current in a wire is
associated with a magnetic field (in the wire itself and in the space surrounding it) and with a
volt-drop in the series resistance of the wire. Both of these effects can be used for current
measurement.
The magnetic field can be measured in many ways:
    Force (the ordinary ammeter is essentially a force measuring device that compares the force
generated by the field with the strength of a spring),
    Magnetoresistance (materials exist whose resistance is a function of the applied magnetic
field),
    Hall Effect (a voltage is generated in a material in proportion to the applied field and the
magnitude of a control current),
    Magnetostriction (materials exist whose physical size is a function of the applied magnetic
field),
    Faraday Effect (rotation of the plane of polarization of light in an optical material).
If the current is ac, there is also the possibility of inductive effects, but we are concerned here
with direct current.
In addition to the magnetic field effects, the volt-drop across a resistor can be measured easily
enough. The question of isolation is not so readily answered, however. A good current
measurement requires that the series resistance be low – and this means that the volt-drop across
it will be small, while the full voltage of the circuit appears as a common-mode voltage in the
measuring system. Typically in the power world, if a transformer is not used, the voltages are so
high that it is necessary to convert the voltage into a signal that can be transmitted via fiber
optics. This conversion requires electronics operating at the full line potential – a possible but not
always simple feat.
For isolation in the magnetic sensors, the most common construction places a magnetic toroid
around the current carrying conductor (thereby forming a single-turn primary winding). This
toroid, and any secondary windings on it, are then easily isolated from the voltage of the
primary.
Table I summarizes the discussion as a taxonomy:

Table I. A taxonomy of current measurement methods
transduction method        measurement effect        isolation
magnetic                   force                     usually results from the fact that
the field extends through all
magnetoresistance
space
Hall Effect
magnetostriction          magnetic field must be high, so
this approach uses optical
isolation
electrical (volt-drop)     voltage                   transformer or fiber optics

For the purposes of the NEPTUNE power system, force-based methods can probably be ruled
out. In most applications of this approach, the force is used simply as a way to obtain the
deflection of a indicator, such as an instrument pointer, so the approach is not applicable. In the
“force-balance” however, while the method is cumbersome, it does have some attractive
attributes. Like a bridge, the deflection is zero at balance, so any aging of mechanical parts will
have minimal effect. And because it works by a feedback method, it is linear over a large
dynamic range.
Magnetoresistance and the Hall effect are both amenable to implementations in which the
measurement element is a “witness” to the magnetic field. The most accurate (and most
common) approach is to place the sensor element in an air gap in a toroid, as shown in Figure 1.
magnetic core
sensing element

air gap

conductor

Figure 1. Current measurement by sensing
the magnetic field

The approach shown in Figure 1 is applicable to the magnetoresistive, Hall, and magnetostrictive
methods of current sensing. It suffers two problems because of the air gap. First, the
measurement is sensitive to external fields, particularly as those caused by currents in nearby
conductors. Second, the accuracy is poor, as the output depends on exactly where in the cross
section of the toroid the sensing element is located. There is also some dependence on the
location of the current-carrying conductor within the magnetic circuit, an effect that is
exacerbated by a large air gap.
A variation of the method dispenses with the magnetic circuit entirely, and uses the sensing
element to “witness” the field in air. This approach has relatively low accuracy, as the output is
normally quite position dependant.
These defects are overcome by the methods in which the integral of the field is measured, rather
than the field at a point near the conductor. The integral is, of course, around a closed loop
encircling the conductor, so that Ampere’s law (I = Hdl) is implemented, and the actual shape
of the closed loop is immaterial. Square loops are used if the sensing is via bulk optics. Circular
ones predominate otherwise. These arrangements are shown in Figure 2.

sensing optics                                    sensing fiber

light path                                         light path

conductor                                         conductor

input fiber    output fiber
input fiber   output fiber

(a) Bulk Optics                                  (b) Fiber Optics

Figure 2. Optical current sensing with the Faraday Effect

The main problem with optical current measurement is it is the rotation of the plane of
polarization of the optical signal that is of interest, and this cannot be measured directly. It is
necessary to convert the rotation to an amplitude, and to cancel out any signal amplitude
variations that are due to other effects (such as connector losses, or temperature). The
components needed to accomplish this are shown in Figure 3.

Light     Polarizer                    Analyzer
Source
Sensing Optics                Detector

Sensor                       D
Input                           Output
Fiber                           Fiber

        

Figure 3. Typical arrangement of optical components in a

Optical current measurement is therefore unavoidably complex, and justified only where the cost
of this complexity can be justified, for example in very high voltage systems.

Isolation
For the optical methods, isolation is achieved simply so long as the measurement electronics are
not close to the conductor. This applies whether the optical path is in free-space, in bulk optics,
or in a fiber, see Figure 2. However, it is doubtful that the complexity of an optical system can be
justified on the NEPTUNE power system, so we will not dwell on the matter.
For the magnetic methods, isolation by a magnetic core should be possible. In the case of
magnetoresistance, Hall, and magnetostrictive transducers, the flux in a magnetic core is
measured in an air-gap in the core, see Figure 1. Provided suitable clearances are arranged,
isolation is simple. However, the problems caused by the air gap warrant our looking further.
In an ac system, current is conventionally scaled by a transformer before it is measured. The
same philosophy is used in HVdc systems, but the transformer is a little different. The primary
current is) arranged to pass through a single conductor in the center of a toroid (as is typical in
power systems. Actually, for dc measurements, there are two toroids. Each toroid has an
excitation winding, supplied from an external source. The excitation windings are arranged in
series opposition. The system direct current affects the magnetization of the two cores in such a
way that the current from the ac source is affected. In some implementations, the excitation
winding is also used as a sense winding. In others, a separate winding is employed. Figure 4
shows the arrangement.
saturable
magnetic core
primary
conductor

excitation and
sensing winding

Figure 4. “Current transformer” for direct current

The technique may have existed in early HVdc schemes, before the invention of the fluxgate
magnetometer in WWII. The principle of operation is the same.

Fault-Current Tolerance
During its expected 30-year life, NEPTUNE may experience as many as a dozen cable faults,
most as a result of fishing. A cable fault results in a large momentary current as the cable
capacitance discharges. The peak value of the current may be as high as a few hundred amps,
many times the normal current being measured, and orders of magnitude higher than the
minimum current that it may be necessary to detect.
Some current measurement methods may be prone to problems because of this current. It is well-
known in ac measurements that toroidal-core devices (such as current transformers) are
susceptible to dc magnetization: this is one of the causes of protection problems in power
systems subject to quasi-static induction during solar wind events. A core that is magnetized
away from the center of its B-H loop will not read accurately, and must be demagnetized by the
application of a large, exponentially-decaying ac excitation.
For NEPTUNE, such magnetization would be possible at all current measurement sites between
the fault and the shore. In view of the cost of accessing such a potentially large number of nodes,
our design for current measurement must either be immune to such problems or it must include
the extra complexity of a demagnetizing scheme.

Leakage Detection
To implement the metering and most of the protection functions of NEPTUNE, it will be
necessary to measure the current at various locations in the node and on shore. However, to
implement the detection of leakage currents on the LV side of the nodes, some other approach
must be used.
At MBARI, the leakage detection approach taken has been to “float” the load circuit, and ground
one side of the supply through a high-value resistor. In order that there should be no place in the
wiring that a fault to ground could exist, each side of the supply is grounded in turn. Since the
grounding resistor is around 1 M, a current as small as 1 A can be detected easily.
A problem with this approach is that there is a transient charging current every time the circuit
ground is moved, and this must be accounted for in the detection system. One way to do this
would be to disable detection until the transient had disappeared. This is how transformer
differential protection is implemented in power systems. Another way would be to compensate
for the current in the detection algorithm. This is what MBARI does.
Another approach would be to ground both sides of the power supply output via a resistor, and to
allow the possibility of a ground fault occurring that resulted in no ground current. Statistically,
this must be unlikely.
An alternative method would be to directly detect the difference in currents on the two sides of
the power supply. That method is considered next.

3.     Review of current comparison methods
An approach called the current comparator, that somewhat resembles the dc current transformer,
has been used for some time in the calibration of ac current transformers. A large current is
passed through two transformers, the one being calibrated and a reference one with nominally
the same ratio. An additional winding is provided on the reference core to drive the flux zero.
(When there is zero flux, the transformer action is described exactly by the turns ratio.) By
arranging the transformers secondaries in series opposition, the flux in the reference core is
nearly zero. The additional drive needed to make it exactly zero is a measure of the errors of the
device under test.
The principles of the fluxgate sensor and the ac current comparator can be combined to produce
a dc current comparator. Such a device has been used to calibrate dc measuring equipment, for
example at the National Research Council of Canada, the Canadian counterpart to NIST.
The minimal configuration of a dc current comparator would consist of a magnetic core with 5
windings. The windings would be two primaries (dc, wound so that the fluxes cancel), an
excitation winding (ac), a sensing winding (ac), and a feedback winding (quasi-dc). A difference
in current in the two primary windings moves the average operating point of the flux up or down
the B-H curve of the material. The nonlinearity that this causes produces second harmonics.
These are detected in a phase-sensitive detector, and used, after filtering, to produce a quasi-dc
feedback to counter the average offset. The ampere-turns necessary to restore the operating point
to symmetry is a measure of the difference current in the two primaries.
More elaborate versions of this arrangement would have multiple cores, often referred to as a
working core and a sensing core, and may employ magnetic shielding. For the purposes of
NEPTUNE, where cost is a factor, the simpler arrangement may be adequate.
The implementation of a low-cost current comparator for NEPTUNE application is discussed in
Appendix A.
4.     Comparison of current measurement methods
Table II compares the current measurement methods discussed above.

Table II. A comparison of current measurement methods
method                 accuracy               frequency       long-term   cost
response        stability
force                  good (<1%)             poor (< 1 Hz)   fair        low (few \$)
magnetoresistance      poor to fair (few %)   good (kHz)      good        low (few \$)
Hall Effect            fair to good (few %)   good (kHz)      fair        low (few \$)
magnetostriction       fair (few %)           good (MHz)      fair        high (k\$)
Faraday Effect         very good (< 1%)       good (MHz)      good        high (k\$)
voltage                very good (< 1%)       good (MHz)      good        high (k\$)

5.     Recommendation
The best method in most respects is the series resistor method, but this method is ruled out by the
cost, which results from dealing with the very large common mode voltage. For NEPTUNE
metering and protection applications, the selection must be made from the low cost entries in
Table II.
The best of the low-cost approaches is the Hall Effect. Since there are several commercial
sources of Hall sensors, this method is recommended for most of the NEPTUNE current
measurements. The question of magnetization by fault current must be borne in mind when
specifying the equipment.
For ground fault detection in the loads, a voltage detection method similar to the one
implemented at MBARI is recommended. Whether it should be balanced or should rely on
switching must be decided by the power group.
Two areas for investigation as student projects at UW are pointed out. It is recommended that a
low-cost implementation of the current comparator be investigated, and a version of the force-
balance. Some ideas are presented in Appendices A and B.
Appendix A
Implementing current comparator electronics

The current comparator is a complex device. Although capable of outstanding performance (the
NRC group in Canada has achieved uncertainties of 1 in 108), cost would seem to be a barrier to
widespread use in NEPTUNE. Figure A-1 shows a block diagram of the electronics needed.

oscillator          ÷2
frequency

magnetic
circuit
output
÷2         excitation                        synch           low-pass
frequency     drive circuit                     detector           filter

feedback
currents
being
compared
Figure A-1. Current comparator block diagram

In this diagram, each of the 5 “connections” to the magnetic circuit represents a winding on the
toroid. The currents being compared may constitute a “single turn” winding, made by passing the
conductor once through the center of the toroid.
Note that an integrator may be included in the feedback loop. It changes the steady-state error,
but not the principle of operation of the system.
Some of the circuitry is similar to that in a high-performance electric field meter developed at
JPL in the late 1980s. Both divide the oscillator frequency by 2 just to ensure signal symmetry.
Both use synchronous switching to achieve the phase-sensitive detector function, both use
filtering to reduce the noise bandwidth and extend the dynamic range. The complexity of the
field meter was such that a fairly large circuit board was required. The current comparator can be
expected to be similar.
To reduce the cost of the current comparator, I wondered if a commercial product could be
“cannibalized.” The obvious choice was the fluxgate magnetometer.
A fluxgate magnetometer, used to measure small magnetic fields and to make electronic
compasses for navigation, uses much of the same hardware as the current comparator. The idea
was first implemented in quantity in WWII as a way for low flying airplanes to detect submerged
submarines, though it may have been around earlier. The biggest difference between a fluxgate
magnetometer and a current comparator is that the sensing winding on the magnetometer is
outside the toroid. It has to be, or the device would be insensitive to external fields. Figure A-2
shows the arrangement, drawn so as to resemble the current comparator.
÷2
oscillator      frequency

magnetic
circuit
output
÷2           excitation                                     synch               low-pass
frequency       drive circuit                                  detector               filter

external
magnetic
field

Figure A-2. Fluxgate magnetometer block diagram

Unfortunately, while fluxgate magnetometers are commercially available, they seem to be
expensive, and to use complex electronics. Although low-cost navigation compasses are
available (for vehicles, for example), they use GMR (giant magnetoresistance effect) technology,
not fluxgate technology.
There may still be a way that some of the current comparator functions can be implemented in a
commercial IC, however. There seems to be considerable overlap between the signal processing
in the current comparator and that in an ordinary FM stereo receiver.
In an FM stereo broadcast, the information for the two stereo channels is transmitted in such a
way that a mono receiver can still perform adequately. Therefore, the mono signal, the sum of
the left and right channels (L+R), is transmitted on a frequency modulated carrier, exactly as it
was before the advent of stereo. In order to separate the channels, the difference (L–R) is
transmitted separately, at a high enough frequency that the mono receiver will not respond to it.
The L–R information is transmitted as an amplitude modulated signal on a 38-kHz subcarrier.
For reasons of transmitter efficiency, the actual subcarrier is suppressed, however. Instead, a
low-level pilot tone at 19 kHz (out of the passband of a mono receiver) is transmitted. This pilot
tone allows stereo receivers to know that the signal is stereo even when there is no modulation
on the subcarrier, and gives phase and frequency information to the AM detector. The spectrum
of an FM signal therefore looks as shown in Figure A-3.

19 kHz            38 kHz
pilot          suppressed
carrier

L+R                                   LR                     LR
lower                   upper
sideband                sideband

0           10               20            30               40              50              60

frequency (kHz, linear scale)

Figure A-3. Stereo signal spectrum
As shown in Figure A-3, after the receiver’s FM demodulator, there is a baseband signal
extending to 15 kHz and containing L+R information. At 19 kHz there is a low-amplitude pilot
tone, unmodulated. Either side of 38 kHz are the sidebands of an AM suppressed-carrier signal
with the L–R information. The amplitude is such that after demodulation, the difference signal
has the same amplitude as the sum signal. The AM signal is demodulated by a synchronous
detector operating at twice the frequency of the pilot tone, and locked to it in phase. By adding
the resulting L–R signal to the L+R signal, a pure L signal is obtained. By subtracting them, a
pure R signal is obtained. These operations are shown in Figure A-4, drawn so as to resemble
Figure A-1 as far as possible.

stereo decoder IC
oscillator             ÷2
76 kHz            frequency

38 kHz

19 kHz

÷2            phase            synch      low-pass
frequency        detector         detector      filter

R
L signal
feedback
L
low-pass         sum &           outputs
filter        dfference   R
input signal

Figure A-4. Stereo decoder IC block diagram

Again, the oscillator frequency is divided by two just to ensure symmetry. The signal is detected
by a synchronous detector, operating at twice the pilot tone frequency, and then passed through a
low-pass filter. In fact, the similarity between the functions in Figures A-1 and A-4 leads one to
wonder whether an FM stereo circuit would function as a current comparator. While I have not
done the experiment, I am inclined to think it would, with the addition of a few components.
There are a few differences between this arrangement and the current comparator. In the stereo
IC, the oscillator phase is locked to the pilot tone, whereas in the current comparator it is free
running. (It may turn out that this is a feature we can use.) The stereo IC contains sum and
difference circuits the current comparator does not need. Perhaps this will not matter. The
spectrum of the current comparator signal should contain nothing in the area of the baseband
L+R signal, so that only the “L–R” signal should appear in the output. In fact, it should appear
identically at both outputs.
Figure A-5 shows a possible use of an FM stereo IC to do the hard work in a current comparator
signal chain.
stereo decoder IC
oscillator              ÷2
76 kHz             frequency

38 kHz

19 kHz

÷2            phase                                        synch          low-pass
frequency        detector                                     detector          filter

R
L signal
feedback

low-pass             sum &                  output
filter            dfference       R
input signal

magnetic
input signal                     circuit

19 kHz
feedback
oscillator               ÷2                    excitation
38 kHz              frequency                drive circuit
currents
being
compared

Figure A-5. Current comparator implemented with stereo decoder IC

The circuit has been largely subsumed into the IC. An external oscillator is still needed, and must
be operated to produce excitation at the pilot tone frequency. The stereo IC locks to that signal
(which will be large relative to the desired signal), and performs the current comparator’s
detection functions.
Difficulties may be experienced due to overload caused by the large size of the “pilot tone.”
Certainly, there should be no problem obtaining lock! Another problem may be the presence in
the signal of significant energy at the third harmonic. Theoretically, the synchronous detection at
2f should eliminate all odd harmonics. An estimate of the comparator spectrum is shown in
Figure A-6.

19 kHz
drive                                                     57 kHz
frequency
third
harmonic
38 kHz
second
harmonic

0              10             20               30              40               50              60

frequency (kHz, linear scale)

Figure A-6. Probable comparator signal spectrum
In addition, it may be necessary to add a driver for the feedback loop to the magnetic circuit.
Nevertheless, the component count is much lower than an conventional implementation would
be, so it seems to be worth the effort of trying it.
Appendix B
Force-balance current measurement

The measurement of current by balancing forces is an ancient technique. Lord Kelvin made one
in around 1882, using a modified weighing-scale and six solenoids, and Lord Raleigh had a
version using coils in the Helmholz arrangement, at around the same time. The method was in
use at the Standards laboratories of both England and the US until the late 1940s as the method
for standardizing the ampere. The principle is illustrated in Figure B1.
balance beam

series-connected
coils

balance weight

Figure B1. Current Balance (after Rayleigh)

In both the Rayleigh and Kelvin versions of the current balance, the force that is balanced is that
due to gravity. In the 1880s this was thought to be pretty constant from place to place, and indeed
the errors that assumption caused were not deemed important for many decades.
A modern variation on the theme could differ in two important ways. First, the current in some
of the coils need not be the measurand, it could be a control current, scaled in some suitable way.
Second the balance – the restoration to zero deflection of some mechanical system – could be
effected by feedback that is measured by some conventional means and becomes the system
output. There would be several advantages: the overall system could be quite robust and
inexpensive, and long-term variations in the stiffness of the deflection mechanism would be
relatively unimportant, affecting the system sensitivity but not the accuracy, and immunity to
large current overload could be by mechanical restraint. Figure B2 shows one possible
implementation.
The proposed system is scarcely more complex than a Hall effect system, and while would have
a very poor frequency response, it would likely confer immunity to surge-current magnetization
problems.
anchor block

deflection        current drive
measurement           circuit

feedback current

restraint
block

current being
measured

Figure B2. Possible implementation of a Current Bridge

The force-balance arrangement shown in Figure B2 is described as a current bridge because it
works by achieving a zero deflection in the position of the middle coil. The position of the coil
can easily be measured by a strain gauge, and doubtless other approaches would also work. The
drive current required to achieve balance is proportional to the measurand, the coefficient of
proportionality including also the number of turns in the various coils and the mutual inductance.
These factors are not likely to change.
The scheme has two major defects. Probably the biggest defect in the measurement is its
expected sensitivity external magnetic fields. Since the magnetics are all air-cored, shielding
must be provided to reduce interference with other such sensors in the area. The other problem is
sensitivity to vibration. It may be possible to make a design that has some vibration immunity.
How serious a problem this will be in NEPTUNE is hard to gauge, as there will likely be no
vibration except during earthquakes and volcanic eruptions
.

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