Computer-Aided Design of Complex Waveguide
Filters for Space Communication Systems
J. V. Morro, C. Bachiller, H. Esteban, V. E. Boria
Instituto de Telecomunicaciones y Aplicaciones Multimedia (iTEAM)
Universidad Politécnica de Valencia
Building 8G, access D, Camino de Vera s/n 46022 Valencia (SPAIN)
Corresponding author: firstname.lastname@example.org
Abstract risk of radiofrequency breakdown decreases. Mo-
reover, if the dielectric posts are circular the ma-
This paper presents a case study of advanced op- nufacturing effort is dramatically reduced com-
timization techniques for the automated design pared to square shapes. However, the accurate
of complex waveguide filters for space appli- modeling of the circular dielectric resonators is far
cation, and a detailed study of the multipactor more complex than for square ones, since circu-
effect in different H-plane waveguide filters: all lar and rectangular geometries must be analyzed
metallic, loaded with dielectric cylinders and together. Other drawback to the use of dielectric
evanescent mode. loading materials in the filters is the increase of
loss level due to the dielectric tangent factor.
Keywords:multipactor, waveguide filters, dielec-
tric cylinders, space communications, computer This paper begins with a case study of advan-
aided design, optimization, aggressive space ced optimization techniques for the automated
mapping, segmentation, hybridization, genetic design of complex waveguide filters for space
algorithms applications. The accurate design of electromag-
netic (EM) structures requires a tradeoff between
1. Introduction accuracy and computation time. When designing
complex structures, the use of a very accurate si-
There are many reasons that lead to develop new mulation tool can be unaffordable. The Aggressive
topologies of high frequency filters for space Space Mapping (ASM) methodologies address this
applications, i.e: reduction of mass and volume, issue. Aggressive Space mapping  can be used
increase of thermal stability for high power appli- to reduce the computational burden by using two
cations, increase of out-of-band rejection, reduc- different simulation tools of different accuracy
tion of manufacturing effort, availability of analy- and efficiency: an efficient but not very accura-
sis and design tools for synthesizing a desired te tool (coarse model) in the optimization space
response and reduction of risk of radiofrequency (OS), and an accurate but not very efficient tool
breakdown (i.e. Multipactor effect , ). (fine model) in the validation space (VS). These
methodologies move the computational burden
Rectangular waveguide H-plane filters are one of to the OS, thus reducing the overall computation
the most popular technologies for implementing time, while the accuracy is still guaranteed by the
satellite communications filters, and many efforts use of the fine model. Although ASM has proved
are being devoted to improve their capabilities. The to be very useful for EM design, there is still much
development of new topologies in this technology research dedicated to improve the robustness
has been historically limited by the availability of and performance of ASM . As an alternative to
CAD tools that allow implementing a filter with a those extensions of ASM, we proposed to improve
required response and several predefined improve- the ASM approach by using a segmentation and
ments in terms of mass, stability or high power ef- hybridization strategy. The speed and robustness
fects (i.e. multipactor). An efficient Computer-Aided of the optimization process can be greatly impro-
Design (CAD) software package requires a fast and ved by decomposing the structure as proposed in
accurate analysis tool for the selected topology  and . Moreover, the design process can still
and a reliable optimization strategy. be improved by using a suitable combination of
several optimization algorithms instead of using
The topologies that are analyzed and designed in a single all-purpose technique such as a genetic
this work are rectangular waveguide H-plane fil- algorithm. In this paper, the completely automa-
ters loaded with cylindrical dielectric posts. When ted CAD tool recently proposed in , which does
introducing these elements in the filter, the mass not require human intervention, is adapted for the
and volume are reduced, the thermal stability and accurate design of several complex waveguide
the out-of-band rejection are increased, and the filters: H-plane coupled cavities filters with and
106 ISSN 1889-8297 / Waves · 2009 · year 1
without tuning elements, and novel designs with
Then, this work presents the results for the multi-
pactor effect in the different topologies of filters
designed, i.e all metallic ones, filters loaded with
circular dielectric posts and evanescent mode fil- Table 4. Types of filters considered in this work.
ters loaded with dielectric posts (see Fig. 1). The
study has been made on the basis of a multimodal
analysis method  that enables the computation ŋ near zero. At each iteration j, the next point is
of the electromagnetic fields inside the filter and found by a quasi-Newton iteration:
the dielectric posts in a very accurate and efficient
way. The results for such electromagnetic fields
have been successfully compared to results ob-
tained with a commercial simulator (Ansoft HFSS 1.2
). Then a comparative study of the multipactor
effect that can appear between the two metallic where x(0) = x* and h(j) solves the linear system:
surfaces of each filter has been performed. In order
to achieve a fair comparison, the study was made B(j)•h(j)=-f(j)
on several filters with the same frequency respon- 1.3
se. The filters loaded with dielectric posts are sma-
ller than all the metallic ones, and some of them B(j) is an approximation to the Jacobian matrix
have also a better out-of-band rejection behavior. and is obtained from B(j-1) using the Broyden up-
Considering the multipactor discharge, the study date .
concludes that the dielectric posts concentrate the
electric field inside them, thus producing a smaller
level of electromagnetic field outside the dielectric 3. Aggressive Space Mapping with
posts than for the all-metallic filters. Following the Segmentation and Hybridization
premise that the multipactor discharge can only
appear between the metallic surfaces of the filter, Segmentation
the dielectric loaded filters can handle more power In  and , a segmentation strategy was pro-
without risk of multipactor breakdown. posed for the design of some filter structures,
such as H-plane coupled cavity filters composed
of N resonant cavities and N+1 coupling win-
2. Aggressive Space Mapping dows. This segmentation technique transforms
Method a slow multidimensional design process into
several efficient and robust design steps, whe-
The original ASM strategy describes the beha- re a small number of parameters are designed
vior of a system by means of two spaces models: at the same time. However, there is the risk that
the optimization space (OS), denoted by Xos, and the coupling among all cavities (not just among
the validation space (VS), denoted by Xem. We adjacent cavities) is not properly taken into ac-
represent the designable model parameters in count. To solve this problem, the segmentation
these spaces by the vectors Xos and Xem, respec- strategy proposed in  adds new steps to the
tively. The objective of the ASM procedure is to original strategy. The resulting new segmenta-
find the optimum point Xem in VS that minimizes tion strategy designs the filter through the steps
the following non-linear function: summarized in Table I.
f(xem )=P(xem )-x*os The Ordinary step designs the parameters rela-
ted to cavity i simulating the i first cavities, and
1.3 using for the rest of parameters of the i-1 first
cavities the values obtained in former iterations.
where X*os is the optimum point in OS and P(xem ) The error function is computed comparing the
is the point in OS that satisfies Rf(xem )=Rc(P(xem)), response of the i first cavities with the ideal res-
Rf and Rc being the vectors with the responses of ponse. The ideal response of the i first cavities is
the fine and coarse models. The ASM procedure obtained using the first i resonators of the proto-
finishes when ||f(xem )|| is below some threshold type composed of impedance inverters and half-
Table 1. Steps of the proposed segmentation design strategy.
Waves · 2009 · year 1 / ISSN 1889-8297 107
wavelength transmission lines. The Coupling the filter, independently of the frequency value.
Both the step adjusts, every three cavities, all the design The input threshold power is also a function of
parameters of the cavities previously designed, the frequency and can be calculated by using
thus achieving the required small changes in the VMFmax, the characteristic impedance Z0
and robust- the values of the parameters due to couplings and the threshold voltage Vmulti that enables
ness can be among cavities. The Central Cavity step designs the multipactor breakdown. This value can be
drastically the dimensions of the central cavity simulating obtained by using Multipactor Calculator 
improved the whole filter structure, and the Final step re- and depends on the type of metallic surface and
when a fines the design taking into account all possible the frequency-gap product.
suitable interactions among cavities.
of optimi- Hybrid Optimization
zation algo- Both the efficiency and robustness of the opti- 1.4
rithms is used mization process can be drastically improved
when a suitable combination of optimization
algorithms is used instead of a single algorithm.
If only a gradient method is used, it may fail to 1.5
reach the optimum if the starting point is far
from it. On the other hand, the use of a robust The minimum of this equation in the whole
method such as the simplex method or genetic frequency band provides the maximum power
algorithm, largely used in circuit design applica- that the device can handle without multipactor
tions, ensures convergence, but at the cost of breakdown.
a low efficiency. So, the design procedure has
been improved using a suitable combination Distribution of electromagnetic field inside
of optimization algorithms in each parameter the filters
extraction phase and also in the optimization This section describes the procedure for obtai-
needed to obtain x*os. Robust non-gradient meth- ning the distribution of electrical field inside the
ods (direct search and simplex) are used at the structures as a function of the frequency.
beginning, and, after some iterations, when we
are close to the minimum, an efficient gradient All-metal cavities filter
algorithms (Broyden-Fletcher-Goldfarb-Shanno The field distribution is obtained from the mul-
(BFGS)) is used to refine the solution. This com- timodal scattering matrices (GSMs) of each one
bination of algorithms has been proved to per- of the segments that compose the structure of
form better than one algorithm alone. The shift the filter. In Fig. 2 the different segments and
from one optimization algorithm to another matrices are shown. GSMs of the step (S2, S4, S6,
one is controlled by the parameter termination S8…) can be calculated following the well-known
tolerance xtol, the termination tolerance of the Mode Matching Method , and the GSMs of
error function (ftol), and the maximum number of the lines (S1, S3, S5, S7…) are obtained through
function evaluations (nFmax) permitted to each analytic expressions. The global GSM is obtained
method. ftol is higher for the first algorithm, and using a new efficient recursively connection
its value is decreased in each subsequent algo- technique proposed in .
rithm. The shift from one algorithm to another,
as well as the rest of the whole design process is Fig. 3 shows the regions where the electric field is
fully automated, so that no human intervention calculated. For each region the field distribution
is needed at all. is obtained by using the following equation :
4. Multipactor breakdown
in waveguide filters
Multipactor effect prediction where
The Hatch and Williams  model has been Mi is the number of modes in the i segment of
used for the study of the multipactor effect insi- the filter. Typically Mi=11.
de the filters. In this model, the maximum input
power without multipactor breakdown is calcu- xi and zi are the coordinates as defined in Fig. 3.
lated by using the electric field distribution insi-
de the structure.
In the model, the “Voltage Magnification Factor”
VMF  is defined as the maximum voltage in
the structure versus the input voltage (Vin) for
all the frequencies. This factor is calculated for all
the points inside the filter and the VMFmax is de-
fined as the maximum of VMFs. During this work
we have found that the maximum of electrical Figure 2. Segments and GSMs matrices of all-
field appeared always in the same point inside metal cavities filter.
108 ISSN 1889-8297 / Waves · 2009 · year 1
am(i) y bm(i) are the modal vectors of incident and
reflected waves between segments i-1 and i as
defined in Fig. 2.
is defined in equation.
bi is the height and ai the width of each section.
Figure 3. Regions for the computation of the electric field.
Filter loaded with dielectric cylinders
Fig. 4 shows the filter with dielectric cylinders
and the different segments in which the filter is
divided. There are three different types of seg-
ments: lines, steps and segment of line loaded
with dielectric cylinder. For each one of them a
different method is used for the computation of
GSMs and the electric field. As in the previous
filter, the GSMs of the steps (S2, S4, S8, S10…) are
calculated by using the Mode Matching Method
 and the GSMs of the lines (S1, S3, S5, S7…) are
computed analytically. For obtaining the GSMs
of the segments loaded with dielectric cylinders Figure 4. Segments and GSMs matrices of the filter loaded with dielectric cylinders.
(S6,S12…) the hybrid mode matching method is
used . Then all the matrices are connected
as described in  in order to get the global
GSM of the filter. For regions i=1, 3, 5, 6a, 7, 9,
11… (see Fig. 5) the electric field is calculated as
described in the previous subsection. The field in
regions 6b, 12b... (see Fig. 5) is obtained by using
the equation , where Jp and H(2) are the Bessel
and second order Hankel functions, and in and cn
are the incident and scattered spectrum coeffi-
cients in cylindrical coordinates.
Figure 5. Regions for the calculation of electric field of the filter loaded with die-
Finally, in regions 6c, 12c… the field is obtained narrow waveguide, while it is propagating in the
by the method described in , as follows: input wider waveguide.
The procedure followed to calculate the field in-
side this filter is similar to the methodology des-
cribed in the two previous sections.
where coefficient sn is: 5. Results
High-Order H-plane waveguide filter for spa-
ce applications at K-Band
The first example under consideration is a con-
ventional H-plane waveguide filter with coupled
cavities for space applications at the K-band. The
ideal transfer function is a standard nine-pole
Chebychev response of 96 MHz bandwidth cen-
1.10 tered at 17.3 GHz.
and r the radius of the dielectric post.
The cavity lengths and coupling aperture widths
Evanescent mode filter loaded with dielectric of the filter have been chosen as design parame-
cylinders ters (see Fig. 8). The input and output wavegui-
In an evanescent mode filter, the fundamental des of the filter, as well as the resonant cavities,
mode TE10, is below the cutoff frequency in the are standard WR-62 waveguides (a=15.799 mm,
Waves · 2009 · year 1 / ISSN 1889-8297 109
Figure 6. Segments and GSMs matrices of evanes-
cent filter with dielectric cylinders
Figure 9. Responses of the H-plane waveguide
filter. Coarse model response at x* versus the fine
model response at xem.
Tunable H-plane waveguide filter for space
In order to test the performance of the design
procedure with more complex structures, two
tunable H-plane coupled cavity filters have been
Figure 7. Regions for calculating the electric field considered. These filters were originally designed
of the evanescent filter with dielectric cylinders. and manufactured in , where the design was
performed manually. The same filters have been
b=7.899 mm). The length of all the coupling win- re-designed with the CAD tool proposed in this
dows has been chosen to be 2 mm. For the design work. The tuning elements are penetrating posts
of this filter, the same modal simulator has been of square cross-section placed at the center of
used both as the coarse and the fine model. This each cavity and each coupling window (see Fig.
modal simulator characterizes the planar discon- 10). As proposed in , the use of these tuning
tinuities using the Method of the Moments (MoM) posts allows the use of a common base structure
according to the traditional Galerkin procedu- for obtaining filters with responses centered at
re . When used as a fine model, the number different frequency bands. The only difference in
of accessible modes, number of basis functions the filters at each frequency band is the penetra-
in the MoM, and number of kernel terms in the tion of the tuning posts.
integral-equation are high enough to obtain very
accurate results. On the other hand, when used as The ideal transfer function is a four-pole stan-
a coarse model, a small number of modes is con- dard Chebychev band-pass response of 300
sidered in order to obtain a faster simulator at the MHz bandwidth centered at 11 GHz and 13 GHz.
expense of a less accuracy. The initial values of The input and output waveguides of the filter, as
the design parameters (x(0)) have been calculated
os well as the resonant cavities, are standard WR-75
according to the method described in . Fig. waveguides (a=19.050 mm, b=9.525 mm). The
9 shows the comparison between the response length of all the coupling windows has been
of the fine model at the final solution in VS (xem) chosen to be 2 mm. The sides of the posts are
and the response of the coarse model at x* . It can
os fixed to 4 mm in the cavities and 2 mm in the
be observed that the desired objective function coupling windows.
has been satisfactorily recovered in the VS. This
solution has required 185 s of CPU time in a 2.4 The design with ASM using segmentation and
GHz Pentium IV PC platform. hybrid optimization required 3 ASM iterations
for both filters under severe convergence cri-
Figure 10. Manufactured tunable filters with tu-
Figure 8. A four cavities H plane rectangular waveguide filter. ning elements. Common base and 11 GHz top.
110 ISSN 1889-8297 / Waves · 2009 · year 1
used for the
Figure 11. Measurements of the manufactured
prototypes with tuning elements.
terion. The design for the filters centered at 11
and 13 GHz required a total CPU times of 49’50’’
and 33’42’’, respectively, in a PC with Pentium
IV processor at 1.7 GHz. Since the fine model is
about 250 times slower than the coarse model,
the total CPU time required for the direct design
of such filters without ASM would be of about Figure 12. H-plane waveguide filter with dielectric
25 hours. This represents an improvement by a resonators.
factor of 30, and clearly proves the advantage of
using ASM for the design of complex waveguide be observed that the desired objective function
devices. The filters have been manufactured in has been satisfactorily recovered in the VS. This
two different pieces, an H-plane base structure solution has required about 2 hours of CPU time
and a separated top cover including all the tun- in a 3 GHz Pentium IV platform. The total length
ing elements. To reduce costs, the same H-plane of the filter (l1+l2+l3+l4+5t) is reduced by al-
base has been used for both filters. The common most 50% when compared with the same filter
H-plane base and the two different tops includ- without dielectric posts, with the correspondent
ing the tuning elements for the filters at 11 and benefit in terms of volume and mass reduction,
13 GHz were manufactured (see Fig. 10) and so critic in satellite communication systems.
measured (see Fig. 11).
Comparative study of multipactor breakdown
H-plane waveguide filter with dielectric reso- The last part of this work evaluates the risk of
nators multipactor breakdown in a set of filters. In or-
The last structure under consideration is an H- der to have a fair comparison the filters should
plane coupled cavities filter with circular dielec- provide the same frequency response, so we
tric posts placed in the middle of each cavity (see designed three different filters: all metal cavities,
Fig. 12(a)). The ideal transfer function is a standard cavities loaded with dielectric cylinders and eva-
four-pole Chebychev response of 300 MHz band- nescent mode loaded with dielectric cylinders
width centered at 11 GHz. The input and output filters, which present the same frequency res-
waveguides of the filter, as well as the resonant ponse shown in Fig. 13.
cavities, are standard WR-75 waveguides (a=19.05
mm, b=9.525 mm). The relative permittivity of the It is important to notice that during the work
dielectric posts is chosen to be 24, and the length we have considered that the multipactor break-
of all the coupling windows have been chosen to down can only appear between two electric
be 2 mm. The remaining dimensions of the struc- plates containing a normal field, so for the cal-
ture (cavity lengths, coupling aperture widths culation of the power that a particular filter can
and radii of the dielectric posts) have been cho- handle we have used the field inside that filter
sen as design parameters. The design procedure but outside the dielectric cylinders. All the die-
described in section III can not be directly applied lectric cylinders used in the filters have a dielec-
to the design of this kind of filters. It is necessary tric permittivity Ɛr=24.
to use a genetic algorithm, since a good start-
ing point can not be easily obtained, as well as All-metal cavities filter
to suppress the segmentation strategy, since the During the study the authors found that the
coupling among cavities is much stronger that in maximum of the electric field was always loca-
all-metallic filters. Again the same simulation tool ted on the central longitudinal axis of the filter,
is used as the coarse and fine model. This simu- and in the middle of the second cavity indepen-
lation tool is described in . It uses a suitable dently of the frequency value. Fig. 14 shows the
combination of an analytical method and a hybrid maximum input and output power without mul-
technique which copes with the different parts of tipactor breakdown as a function of the frequen-
the filter structure. Fig. 12(b) shows the compari- cy. It is interesting that inside the pass band the
son between the response of the fine model at amount of input and output power that the filter
xem and the ideal response of the filter. It can can handle is the same, around 2500 W, and that
Waves · 2009 · year 1 / ISSN 1889-8297 111
Figure 13. Scattering parameters S11 and S21 of
the H-plane filters under study. Figure 15. Maximum output power without mul-
tipactor breakdown for the cavities filter loaded
with dielectric cylinders, loss tangents tgδ=0,
Figure 14. Maximum input and output power that
the all-metal cavities filter can handle without multi-
pactor breakdown as a function of frequency.
Figure 16. Maximum output power without mul-
outside this band the power is not limited by the tipactor breakdown for the evanescent mode filter
multipactor risk but by the electromagnetic res- loaded with dielectric cylinders, loss tangents
ponse of the filter. tgδ=0, tgδ=10-3, tgδ=10-4
Filter loaded with dielectric cylinders ller than the all-metal one, which makes that the
In this filter the electric field was concentrated electric field is more concentrated in the filter
inside the dielectric cylinders, specifically there and that the risk of multipactor breakdown in-
was a maximum of field inside the second cylin- creases. Nevertheless, when loading this filter
der, being the maximum of field outside the die- with dielectric cylinders, the output power wi-
lectric cylinders considerably lower than in the thout multipactor breakdown (see Fig. 16) can be
case of the all-metal filter. Nevertheless, loading higher than in the all-metal filter, since the field
the filters with a dielectric material can introdu- trends to concentrate inside the dielectric cylin-
ce losses in the filter response that can affect the ders. The figure also shows the output power for
output power that the filter can handle. Fig. 15 different dielectric materials, with different loss
shows the output power that the filter can han- tangent factors, concluding that a good quality
dle without the risk of multipactor breakdown factor dielectric (tgδ=10-4) can provide a power
for dielectric materials with different loss factors. response similar to the ideal one, while a worse
Even with a loss tangent of 10-3 this filter can dielectric reduces this power handling but main-
handle the double of power than the all-metal tains it above the all-metal one.
one. Moreover, with a rather high quality factor
(tgδ=10-4) the behavior of the filter does not di- Comparison of results
vert from the ideal (lossless) case. It is interesting to show a comparative study of the
three filters considering three relevant items: size,
Evanescent mode filter loaded with dielectric maximum field inside the filters and maximum
cylinders output power without multipactor risk. Table II
The evanescent mode filter is substantially sma- lists the total length of the three filters. The eva-
nescent filter presents a reduction in size of about
50% comparing to the all-metal case, and the cavi-
ties filter with dielectric cylinders is about 60% of
Table 2. Lenghts of the three filters. the original size. The following table presents the
maximum value of the electric field inside the fil-
ters, for both cases: inside the dielectric cylinders
and outside them. When having filters loaded with
dielectric materials, the maximum of field is always
located inside the dielectric cylinders. This pheno-
menon enables the increase of power without risk
Table 3. Maximum field (tgδ =1e-4) of multipactor as shown in Fig. 17. Fig. 17 presents
112 ISSN 1889-8297 / Waves · 2009 · year 1
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Waves · 2009 · year 1 / ISSN 1889-8297 113
Theory and Techniques, vol. 55, no. 9, pp 1880- Valencia as an assistant lecturer. She is secretary of
1886, September 2007 the iTEAM Institute of Multimedia Technology and
 H. Esteban,“Análisis de Problemas Arbitrarios Communications of the Polytechnic University of
de Dispersión Electromagnética Mediante Valencia. She is teaching signal and systems theory
Métodos Híbridos” PhD Thesis, Departamento
, and microwaves. She has participated in several
de Comunicaciones, Universidad Politécnica teaching innovation projects as the development
de Valencia, Spain 2002. of GUIs for teaching electromagnetic phenomena.
 G. Gerini, M. Guglielmi, and G. Lastoria, She is now working in her Ph.D. Thesis on electro-
“Efficient integral equation formulations for magnetism and radiofrequency circuits.
admittance or impedance representation of
planar waveguide junctions,” IEEE MTT-Symp.
Dig., vol. III, pp. 1747-1750, 1998. Héctor Esteban González
 P. Soto, J. Gómez, A. Bergner, V. E. Boria, and R. received the degree in Te-
Chismol, “Automated design of waveguide fil- lecommunications Engi-
ters using space mapping optimization,” in Proc. neering from the Polytech-
of 3rd Eu. Conf. on Numerical Meth. in Electro- nic University of Valencia
magnetism, Poitiers, March 2000, pp. 228-229. (UPV), Spain, in 1996, and
 V. E. Boria, M. Guglielmi, and P. Arcioni, “Com- the Ph.D. degree in 2002.
puter-aided design of inductively coupled He collaborated with the
rectangular waveguide filters including tu- Joint Research Centre, Eu-
ning elements,” Int. J. of RF and Microwaves ropean Commission, Ispra, Italy. In 1997, he was
Computer-Aided Engineering, vol. 8, no. 3, pp. with the European Topic Centre on Soil (Euro-
226-236, May 1998. pean Environment Agency). He rejoined the UPV
 C. Bachiller, H. Esteban, V. E. Boria, J. Morro, L. in 1998. His research interests include methods
Rogla, M. Taroncher, and A. Belenguer,“Efficient for the full- wave analysis of open-space and gui-
CAD tool fo direct-coupled-cavities filters with ded multiple scattering problems, CAD design of
dielectric resonators,” in 2005 IEEE AP-S Int. microwave devices, electromagnetic characte-
Symp. Dig., Washington D.C., June 2005. rization of dielectric and magnetic bodies, and
the acceleration of electromagnetic analysis me-
thods using the wavelets and the FMM.
Vicente E. Boria Esbert
José Vicente Morro received the Ingeniero de
received the Telecom- Telecomunicación and the
munications Engineering Doctor Ingeniero de Te-
degree from the Universi- lecomunicación degrees
dad Politécnica de Valen- from the Universidad Po-
cia (UPV), Valencia, Spain, litécnica de Valencia, Va-
in 2001, and is currently lencia, Spain, in 1993 and
pursuing his Ph.D. degree 1997, respectively. In 1993
at UPV. In 2001, he beca- he joined the Departamento de Comunicacio-
me a Research Fellow with the Departamento nes, Universidad Politécnica de Valencia, where
de Comunicaciones, UPV. In 2003 he joined the he is Full Professor (since 2003). In 1995 and 1996
Signal Theory and Communications Division, he was held a Spanish Trainee position with the
Universidad Miguel Hernández, where he was a European Space research and Technology Centre
Lecturer. In 2005, he rejoined the Departamento (ESTEC)-European Space Agency (ESA), Noordwi-
de Comunicaciones, UPV, as an Lecturer. His cu- jk, the Netherlands. Since 2003, he has served on
rrent interests include CAD design of microwave the Editorial Boards of the IEEE Transactions on
devices and EM optimization methods. Microwave Theory and Techniques. He is also
member of the Technical Committees of the
IEEE-MTT International Microwave Symposium
Carmen Bachiller and of the European Microwave Conference. His
received her degree in current research interests include numerical me-
Communication Enginee- thods for the analysis of waveguide and scatte-
ring from the Polytechnic ring structures, automated design of waveguide
University of Valencia in components, radiating systems, measurement
1996. She worked from techniques, and power effects in passive wave-
1997 to 2001 in the com- guide systems.
pany ETRA I+D, S.A as a
Project Engineer in re-
search and development on automatic traffic
control, public transport management and public
information systems using telecommunication
technology. In 2001 she joined the Communica-
tion Department of the Polytechnic University of
114 ISSN 1889-8297 / Waves · 2009 · year 1