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Dynamic CMOS Circuits

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```									Chapter 6

Dynamic CMOS Circuits
Boonchuay Supmonchai
Integrated Design Application Research (IDAR) Laboratory

August 15, 2004; Revised - July 4, 2005

B.Supmonchai

Goals of This Chapter


In-depth discussion of CMOS logic families
 Static and Dynamic  Pass-Transistor
 Nonratioed and Ratioed Logic



Optimizing gate metrics
 Area, Speed, Energy or Robustness



High Performance circuit-design techniques

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Dynamic CMOS


In static circuits at every point in time (except when switching) the output is connected to either GND or VDD via a low resistance path.
 fan-in of N requires 2N devices



Dynamic circuits rely on the temporary storage of signal values on the capacitance of high impedance nodes.
 requires only N + 2 transistors  takes a sequence of precharge and conditional

evaluation phases to realize logic functions
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Dynamic Gate
CLK In1 In2 In3 CLK
Mp

CLK Out CL

Mp

on off

1 Out !((A&B)|C) C

PDN

A

B
Me

CLK

Me

off on

Two phase operation Precharge (CLK = 0) Evaluate (CLK = 1)
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Conditions on Output


Once the output of a dynamic gate is discharged, it cannot be charged again until the next precharge operation. Inputs to the gate can make at most one transition during evaluation. Output can be in high impedance state during and after evaluation (PDN off), state is stored on CL
 This behavior is fundamentally different than the static





counterpart that always has a low resistance path between the output and one of the power rails
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Properties of Dynamic Gates


Number of transistors is N + 2 (versus 2N for static complementary CMOS)
 Logic function is implemented by the PDN only  Should be smaller in area than static complementary CMOS



Full swing outputs (VOL = GND and VOH = VDD) Nonratioed - sizing of the devices is not important for proper functioning (only for performance) Low noise margin (NML)
 PDN starts to work as soon as the input signals exceed VTn, so





set VM, VIH and VIL all equal to VTn
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Properties of Dynamic Gates II


Faster switching speeds
 Reduced load capacitance due to lower number of transistors

per gate (Cint) so a reduced logical effort
 Reduced load capacitance due to smaller fan-out (Cext)  No Isc, so all the current provided by PDN goes into

discharging CL
 Ignoring the influence of precharge time on the switching

speed of the gate, tpLH = 0 but the presence of the evaluation transistor slows down the tpHL


Needs a precharge/evaluate clock

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Properties of Dynamic Gates III


Power dissipation should be better than CMOS
 Consumes only dynamic power – no short circuit power

consumption since the pull-up path is not on when evaluating
 Lower CL- both Cint (since there are fewer transistors

connected to the drain output) and Cext (since there the output load is one per connected gate, not two)
 No glitches - By construction can have at most one transition

per cycle


However overall power dissipation is usually higher than static CMOS due to
 higher transition probabilities  extra load on CLK

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Dynamic Behavior
CLK
Out In1
1.5 2.5

Evaluate

In2
In3 In4
0.5

In & CLK

Out

Precharge
1

-0.5
0 0.5

CLK

all data inputs set to 1

Time (ns)

#Trs 6
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VOH 2.5V

VOL 0V

VM VTn

NMH 2.5-VTn

NML VTn

tpHL 110ps

tpLH 0ns

tp 83ps
9

Dynamic CMOS Gates

B.Supmonchai

Notes on Dynamic Behavior


The precharge time is determined by the time it takes to charge CL through the PMOS precharge transistor.
 Often, the overall digital system can be designed in such a way

that the precharge time coincides with other system functions (e.g., precharge of a FU can coincide with instruction decode).


The duration of the precharge cycle can be adjusted by changing the size of the PMOS precharge transistor.
But making it too large increases the gate’s Cint as well as increasing the capacitive load on the clock.



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Gate Parameters are Time Independent


The amount by which the output voltage drops is a strong function of the input voltage and the available evaluation time.
 Noise needed to corrupt the signal has to be larger if the

evaluation time is short – i.e., the switching threshold is truly time independent.
2.5

CLK
Vout (VG=0.45)

Voltage (V)

1.5

Vout (VG=0.5)
0.5

Vout (VG=0.55)
-0.5 0 20 40 60 80 100

Time (ns)
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Power Consumption of Dynamic Gate
CLK In1 In2 In3 CLK
Mp

Out CL PDN
Eliminates Static power Consumption

Me

Power only dissipated when previous Out = 0
But what about clock power impact?
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Dynamic PC is Data Dependent
Dynamic 2-input NOR Gate
A 0 0 1 1 B 0 1 0 1 Out 1 0 0 0

Assume signal probabilities PA=1 = 1/2 PB=1 = 1/2
Then transition probability P01 = Pout=0 x Pout=1

= 3/4 x 1 = 3/4 Switching activity can be higher in dynamic gates! P01 = Pout=0
(static NOR gate P01 = 3/16)
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Issues in Dynamic Design 1:
CLK
4

Charge Leakage

CLK

Mp

3

Out
1

A=0
2

CL
Me

CLK

VOut
Precharge

Evaluate

Leakage sources

Minimum clock rate of a few kHz
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Source of Charge Leakage


Charge stored on CL will leak away with time (input in low state during evaluation)
Dominant leakage sources are reverse-biased diode (1) and the sub-threshold leakage (2) of the NMOS pulldown device.





PMOS precharge device also contributes some leakage due to reverse bias diode (3) and subthreshold conduction (4) that, to some extent, offsets the leakage due to the pull down paths. Requires a minimum clock rate
 Not good for low performance products such as watches (or



when there are conditional clocks)
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Impact of Charge Leakage


Output settles to an intermediate voltage determined by a resistive divider of the pull-up and pull-down networks
 Once the output drops below the switching threshold of the

fan-out logic gate, the output is interpreted as a low voltage.
CLK
2.5 Voltage (V)

Out
1.5
0.5

-0.5
0 20 40

Time (ms)
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A Solution to Charge Leakage


Keeper compensates for the charge lost due to the pulldown leakage paths.
Keeper
CLK A B
CLK

Mp Mkp
Out

Same approach as level restorer logic

CL

State Me Precharge Evaluate

PDN Irr.
OFF ON

Out VDD
VDD VDD  0

Mkp ON
ON ON  OFF

If PDN is on, there is a fight between the PDN and the PUN - circuit must be ratioed so that PDN wins, eventually
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Issues in Dynamic Design 2:
CLK
A
Mp

Charge Sharing

Out CL

B=0
CLK
Me

Ca
Cb

Charge stored originally on CL is redistributed (shared) over CL and CA leading to static power consumption by downstream gates and possible circuit malfunction.

When Vout = - VDD (Ca / (Ca + CL )) the drop in Vout is large enough to be below the switching threshold of the gate it drives causing a malfunction.
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Charge Sharing Example
What is the worst case voltage drop on y? (Assume all inputs are low during precharge and that all internal nodes are initially at 0V.)
CLK
A !A

y=ABC

Load inverter

a
Ca=15fF

Cy=50fF

B

c

!B

b
B !B

d

Cb=15fF

Cc=15fF

!C

C

Cd=10fF

CLK

Vout = - VDD [(Ca + Cc)/((Ca + Cc) + Cy)]
= - 2.5V*(30/(30+50)) = -0.94V
Dynamic CMOS Gates 19

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Notes on Charge Sharing Example


Output stays high for 4 out of 8 cases (!A B C, !A !B !C, A !B C, and A B !C) Worst case is obtained by exposing the maximum amount of internal capacitance to the output node during evaluation.
 This happens when !A B C or A !B C





∆V = -0.94 V so the output drops to 2.5 - 0.94 = 1.56 V which is below the switching threshold of the Load inverter.
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B.Supmonchai

Solution to Charge Redistribution
CLK
Mp Mkp

CLK

Out A B
CLK
Me

Precharge internal nodes using a clock-driven transistor (at the cost of increased area and power)
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Issues in Dynamic Design 3:


Backgate Coupling

Susceptible to crosstalk due to 1) high impedance of the output node and 2) capacitive coupling
CLK
Mp

Out1 =1 CL1

M6 M5

Out2 =0 A=0
B=0
CLK
M1
M2 M4

CL2 In

M3

Me

Static NAND

Dynamic NAND

Out2 capacitively couples with Out1 through the gate-source and gate-drain capacitances of M4
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Backgate Coupling Effect


Capacitive coupling means Out1 drops significantly so Out2 does not go all the way to ground
3

2

Out1
1

CLK

0

In
0 2

Out2

-1 4 6

Time (ns)
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Notes on Backgate Coupling Effect


The high impedance of the output node makes the circuit very sensitive to crosstalk effects.
 A wire routed over or next to a dynamic node may

couple capacitively and destroy the state of the floating node.


Due to capacitive backgate coupling between the internal and output node of the static gate and the output of the dynamic gate, Out1 voltage is reduced. Out1 overshoots VDD (2.5V) due to clock feedthrough
Dynamic CMOS Gates 24



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Issues in Dynamic Design 4:


Clock Feedthrough

A special case of capacitive coupling between the clock input of the precharge transistor and the dynamic output node
Coupling between Out and CLK input of the precharge device due to the gate- drain capacitance. So voltage of Out can rise above VDD. The fast rising (and falling edges) of the clock couple to Out.
Dynamic CMOS Gates 25

CLK
A

Mp

Out CL

B

CLK

Me

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Clock Feedthrough Example
CLK

Clock feedthrough
Out
2.5

In1 In2 In3
In4
CLK
-0.5 0 0.5 1 1.5

0.5

In & CLK Out

Clock feedthrough

Time (ns)

Signal levels can rise enough above VDD that the normally reversebiased junction diodes become forward-biased causing electrons to be injected into the substrate.
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Cascading Dynamic Gates
V
CLK
Mp CLK Mp

CLK

Out1

Out2

In
CLK
Me CLK Me

In
Out1

VTn V t

Out2

Only a single 0  1 transition allowed at the inputs during the evaluation period!
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Domino Logic
CLK In1 In2 In3
CLK
Mp
11 10

CLK
Out1
00 01

Mp Mkp Out2

PDN

In2 In3 CLK

PDN

Me

Me

Assume all inputs to the Domino gate are initially zero
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Why Domino?
CLK

In1 Ini Inj
CLK

PDN

Ini Inj

PDN

Ini Inj

PDN

Ini Inj

PDN

Like falling dominos!
Dynamic CMOS Gates 29

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Notes on Dominic Logic


Ensures all inputs to the Domino gate are set to 0 at the end of the precharge period. Hence, the only possible transition during evaluation is 0 to 1 Additional advantage is that the fan-out of the gate is driven by a static inverter with a low-impedance output that increases the noise immunity. The buffer also reduces the capacitance of the dynamic output node by separating internal and load capacitances. Finally, the inverter can be used to drive a bleeder to combat leakage and charge redistribution.
Dynamic CMOS Gates 30







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Domino Manchester Carry Chain
CLK

3

P0

3

P1

3

P2

3

P3

3 Ci,4
1

Ci,0
CLK

4 5 G0 6

3
4 G1

2
3 G2

1
2 G3

5

4

3

2

!(G0 + P0 Ci,0)

!(G1 + P1G0 + P1P0 Ci,0)

Automatically forms all the intermediate carries
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Domino Comparator
CLK

A3

A2

A1

A0

Out

B3

B2

B1

B0

Don’t need isolation NMOS in the pull-down, since the PDN is forced off during precharge.
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Properties of Domino Logic


Only non-inverting logic can be implemented, fixes include
 can reorganize the logic using Boolean transformations  use differential logic (dual rail)  use np-CMOS (zipper)



Very high speed
 tpHL = 0, only Low-High transitions allow  static inverter can be optimized to match fan-out (separation of

fan-in and fan-out capacitances)
 Input capacitances reduced - smaller logical effort
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Differential (Dual Rail) Domino


Solve problem of non-inverting logic
off
CLK
Mp Mkp

on
Mkp Mp

CLK

Out = AB 1 A

0
!A B CLK
Me

1
!B

0 !Out = !(AB)

AND/NAND Due to its high-performance, differential domino is very popular and is used in several commercial microprocessors!
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Notes on Differential Domino


The inputs and their complements come from other differential DR gates and thus all inputs are low during precharge and make a conditional transition from 0 to 1.
Expensive - but can implement any arbitrary function.

 

Use significant power since they have a guaranteed transition every single clock cycle (regardless of signal statistics, since either Out or !Out will transit from 0 to 1). Nonratioed (even though it has a cross-coupled PMOS pair)



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np-CMOS (Zipper)
CLK
Mp

!CLK
11 10

Me

Out1

In1 In2 In3 CLK

PDN

In4 In5 !CLK

PUN
00 01

Me

Mp

Out2 (to PDN)

In4 and In5 must be from PDN

Only 0  1 transitions allowed at inputs of PDN Only 1  0 transitions allowed at inputs of PUN
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NORA (No Race)
CLK In1 In2 In3 CLK
Mp

!CLK
11 10

Me

Out1

PDN

In4 In5 !CLK
to other PDN’s

PUN
00 01

Me

Mp

Out2 (to PDN)

to other PUN’s

Very sensitive to Noise!
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Note on np-CMOS and NORA
 

DEC alpha uses np-CMOS logic (Dobberpuhl) Have to size the PUN’s to equalize the delay to that of the PDN’s Really dense layouts and very high speed (20% faster than domino with the correct sizing) Reduced noise margin (as with any dynamic gate)
 More sensitive to noise







Increase complexity
 Have two clock signals to generate and route - CLK and !CLK
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np-CMOS Adder Circuit
!CLK
1x 0x

CLK !C1 !A1 !B1

1x

Sum1

!A1

!B1

!B1 !A1

!A1
0  xC2

!B1

!C1
CLK !CLK

!CLK

CLK
0x

1  x !C1

A0
B0 CLK B0

A0

B0
1x

C0

A0

B0 A0 C0

C0
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!CLK
Dynamic CMOS Gates

0x

!Sum0

39

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How to Choose a Logic Style


Must consider ease of design, robustness (noise immunity), area, speed, power, system clocking requirements, fan-out, functionality, ease of testing
4-input NAND

Style Comp Static CPL* domino DCVSL*
* Dual Rail


# Trans 8 12 + 2 6+2 10

Ease 1 2 4 3

Ratioed? Delay Power no no no yes 3 4 2 1 1 3 2 + clk 4

Current trend is towards an increased use of complementary static CMOS: design support through DA tools, robust, more amenable to voltage scaling.
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